High-speed serial data transceiver and related methods

ABSTRACT

A high-speed serial data transceiver includes multiple receivers and transmitters for receiving and transmitting multiple analog, serial data signals at multi-gigabit-per-second data rates. Each receiver includes a timing recovery system for tracking a phase and a frequency of the serial data signal associated with the receiver. The timing recovery system includes a phase interpolator responsive to phase control signals and a set of reference signals having different predetermined phases. The phase interpolator derives a sampling signal, having an interpolated phase, to sample the serial data signal. The timing recovery system in each receiver independently phase-aligns and frequency synchronizes the sampling signal to the serial data signal associated with the receiver. A receiver can include multiple paths for sampling a received, serial data signal in accordance with multiple time-staggered sampling signals, each having an interpolated phase.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.11/446,144, filed Jun. 5, 2006, now allowed, which is a continuation ofU.S. patent application Ser. No. 09/844,441, filed Apr. 30, 2001, nowU.S. Pat. No. 7,058,150, which claims priority to U.S. ProvisionalApplication No. 60/200,813, filed Apr. 28, 2000, all of which areincorporated herein by reference in their entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to data transceivers.

2. Background Art

A communication device including a transmitter and a receiver is knownas a transceiver. Known transceivers can transmit and receive datasignals. There are demands on such transceivers to transmit and receivesuch data signals with low error rates and at ever increasing datarates, to reduce power dissipation, cost, and size. Therefore, there isa general need for a transceiver capable of satisfying such demands.

It is desirable to integrate transceiver circuits on an integratedcircuit (IC) chip to reduce size and power dissipation of thetransceiver. The circuits on the IC chip typically operate in accordancewith timing signals. However, oscillators used to generate such timingsignals have disadvantages, including typically large sizes, high powerdissipation, and deleterious electromagnetic radiative properties (thatis, the oscillators tend to radiate electromagnetic interference acrossthe IC chip). Also, oscillators used in communication devices often needto be tunable in both phase and frequency and in response to rapidlychanging signals. This requires complex oscillator circuitry. Moreover,multiple oscillators on a common IC chip are subjected to undesiredphenomena, such as phase and/or frequency injection locking, whereby oneoscillator can deleteriously influence the operation of anotheroscillator.

Therefore, there is a general need to integrate transceiver circuits onan IC chip. There is a related need to reduce the number and complexityof oscillators constructed on the IC chip, to thereby avoid orsubstantially reduce all of the above-mentioned disadvantages associatedwith such oscillators.

To reliably process a received data signal, a receiver typically needsto match its operating characteristics with the characteristics of thereceived data signal. For example, in the case of baseband datatransmissions, the receiver can derive a sampling signal, and then usethe sampling signal to sample the received data signal at sample timesthat produce optimal data recovery. In this way, data recovery errorscan be minimized.

Precision timing control techniques are required to achieve and maintainsuch optimal sampling times, especially when the received data signalshave high data rates, such as multi-gigabit-per-second data rates. Suchtiming control includes control of the phase and frequency of a samplingsignal used to sample the received data signal.

As the received data signal rate increases into themulti-gigabit-per-second range, the difficulty in effectivelycontrolling sampling processes in the receiver (such as controllingphase and frequency characteristics of the sampling signal)correspondingly increases. For example, semiconductor circuits, such ascomplementary metal oxide semiconductor (CMOS) circuits, are oftenunable to operate at sufficiently high frequencies to optimally controlthe sampling processes. For example, it becomes increasingly difficultat such high received signal data rates to provide sufficiently shorttime delays usable for controlling sampling phases of the samplingsignal.

Accordingly, there is a need for systems and techniques in a datareceiver that provide effective sampling of high data rate signals.There is a related need to reduce the number of circuit componentsrequired to provide such effective data signal sampling, therebyreducing cost, size, and power dissipation in the data receiver.

BRIEF SUMMARY OF THE INVENTION I. Phase Interpolator

The present invention is directed to a phase interpolation system. Thephase interpolation system includes a stage controller adapted toproduce a plurality of stage control signals, and a plurality ofreference stages that are each adapted to convert one of a plurality ofreference signals into a corresponding component signal. Each referencestage performs this conversion in response to a respective one of thestage control signals. Each of the component signals has a distinctphase that is determined by the corresponding reference signal phase.

The phase interpolation system also includes a combining node that isadapted to combine (e.g., sum) the component signals into an outputsignal having an interpolated phase.

Each of the plurality of reference stages may include a conversionmodule and one or more scaling modules. The conversion module is adaptedto convert the corresponding reference signal into the correspondingcomponent signal according to a scaling factor. The one or more scalingmodules are adapted to adjust the scaling factor in response to a valueof the corresponding stage control signal.

Each of the stage control signals may include a plurality of binarycontrol subsignals. In this embodiment, the value of each stage controlsignal is the sum of the corresponding binary control signals. Each ofthese subsignals may be received by one of a plurality of scalingmodules. As a result, the scaling factor of the respective referencestage increases with the value of the corresponding stage controlsignal.

In a specific implementation, four reference stages are each adapted toconvert one of four reference signals into a corresponding componentsignal in response to a respective one of the stage control signals.These four reference signals each have one of four phases that areseparated at substantially 90 degrees intervals.

The conversion module of each reference stage may include atransconductance device, such as a field effect transistor (FET).

The output signal as well as each of the reference and component signalsmay be differential signals.

The stage controller may be a phase control signal rotator adapted toadjust the plurality stage control signals such that the output signalis phase aligned with a serial data signal.

Without the use of conventional techniques, such as time-delays, thephase interpolator advantageously provides output signal phases thatspan a complete rotation of 360 degrees.

II. Timing Recovery System

A receiver of the present invention includes a timing recovery system torecover timing information from a received serial data signal. Thereceiver uses such recovered timing information to compensate forfrequency and phase offsets that can occur between the received serialdata signal and a receiver sampling signal used to sample the serialdata signal. The timing recovery module of the present inventionrecovers/extracts phase and frequency information from the receivedserial data signal. The timing recovery module derives the samplingsignal using the phase and frequency information. The timing recoverymodule phase aligns and frequency synchronizes the sampling signal withthe serial data signal to enable the receiver to optimally sample theserial data signal.

The timing recovery system of the present invention includes a phaseinterpolator. The phase interpolator derives a sampling signal having aninterpolated phase in response to 1) phase control inputs derived by thetiming recovery system, and 2) a set of reference signals derived from amaster timing signal. The timing recovery system causes the interpolatorto align the interpolated phase of the sampling signal with the serialdata signal phase. In addition, the timing recovery system can cause theinterpolator to rotate the interpolated phase of the sampling signal ata controlled rate to synchronize the sampling signal frequency to theserial data signal frequency.

The present invention advantageously simplifies a master oscillator usedto generate the master timing signal (mentioned above) because the phaseinterpolator, not the oscillator, tunes the phase and frequency of thesampling signal. In other words, the master oscillator need not includecomplex phase and frequency tuning circuitry, since the need for suchfunctionality is met using the timing recovery system. Additionally,multiple, independent timing recovery systems can operate off of asingle, common master timing signal, and thus, a single masteroscillator. This advantageously reduces to one the number of masteroscillators required in a multiple receiver (that is, channel)environment on an IC chip. In such a multiple receiver environment, eachof the multiple independent timing recovery systems (and interpolators)can be associated with each one of the multiple receivers. Each timingrecovery system can track the phase and frequency of an associated oneof multiple receive data signals, thus obviating the need for more thanone oscillator.

In one embodiment, the present invention is directed to a system forrecovering timing information from a serial data signal. The systemcomprises a phase interpolator adapted to produce a timing signal havingan interpolated phase responsive to a plurality of phase controlsignals. The system further comprises a phase controller adapted toderive a rotator control signal based on a phase offset between thereceived data signal and the timing signal. The system further comprisesa phase control signal rotator adapted to rotate the plurality of phasecontrol signals and correspondingly the interpolated phase of the timingsignal in response to the rotator control signal. The phase controlleris adapted to cause the phase control signal rotator to rotate theplurality of phase control signals and correspondingly the interpolatedphase of the timing signal in a direction to reduce the phase offsetbetween the received data signal and the timing signal. The rotatorcontrol signal is one of a phase-advance, a phase-retard, and aphase-hold signal. The phase control signal rotator rotates theplurality of phase controls signals in a first direction to advance theinterpolated phase of the timing signal in response to the phase-advancesignal, rotates the plurality of phase controls signals in a seconddirection to retard the interpolated phase in response to thephase-retard signal, prevents the plurality of phase control signals andcorrespondingly the interpolated phase from rotating in response to thephase-hold signal.

In another embodiment, the present invention is directed to a method ofrecovering timing information from a serial data signal. The methodcomprises deriving a timing signal having an interpolated phase inresponse to a plurality of phase control signals, deriving a rotatorcontrol signal based on a phase offset between the received data signaland the timing signal, and rotating the plurality of phase controlsignals and correspondingly the interpolated phase of the timing signalin response to the rotator control signal.

In still another embodiment, the present invention is directed to asystem for recovering timing information from a serial data signal. Thesystem comprises a phase interpolator adapted to derive a samplingsignal having an interpolated phase based on a plurality of controlsignals. The system further comprises a controller coupled to the phaseinterpolator. The controller includes a phase error processor adapted toderive an estimate of a frequency offset between the sampling signal andthe serial data signal. The controller causes the phase interpolator torotate the interpolated phase of the sampling signal at a ratecorresponding to the frequency offset so as to reduce the frequencyoffset between the sampling signal and the serial data signal.

In yet another embodiment, the present invention is directed to a methodof recovering timing information from a serial data signal. The methodcomprises deriving a sampling signal having an interpolated phase,estimating a frequency offset between the sampling signal and the serialdata signal, and rotating the interpolated phase of the sampling signalat a rate corresponding to the frequency offset, thereby reducing thefrequency offset between the sampling signal and the serial data signal.The method also comprises repetitively rotating the interpolated phaseof the sampling signal through a range of phases spanning 360° at therate corresponding to the frequency offset. The method also comprisesrotating the interpolated phase of the sampling signal in a direction ofincreasing phase to decrease a frequency of the sampling signal when thefrequency of the sampling signal is greater than a frequency of theserial data signal, and rotating the interpolated phase of the samplingsignal in a direction of decreasing phase to increase a frequency of thesampling signal when the frequency of the sampling signal is less thanthe frequency of the serial data signal.

III. High-Speed Serial Data Transceiver

The present invention provides a multiple-receiver transceiver (alsoreferred to as a multi-channel transceiver), on an IC chip. This is alsoreferred to herein as a multi-channel communication device, on an ICchip. The communication device advantageously includes only a singlemaster timing generator (that is, oscillator module), to reduce powerconsumption, size, part count and complexity, and avoid problemsassociated with multiple oscillator architectures, such as thosedescribed above. Each receiver in the communication device can process(that is, recover data from) a respective received, analog serial datasignal having a multi-gigabit-per-second data rate. Each receiver isassociated with an independently operating timing recovery system,including a phase interpolator, for phase and frequency tracking therespective received, analog serial data signal.

In an embodiment, the present invention is directed to a communicationdevice on an IC chip. The communication device comprises a master signalgenerator adapted to generate a master timing signal, and a receive-laneadapted to receive an analog serial data signal. The receive-laneincludes a sampling signal generator adapted to generate multipletime-staggered sampling signals based on the master timing signal, andmultiple data paths each adapted to sample the serial data signal inaccordance with a corresponding one of the time-staggered samplingsignals. The multiple data paths thereby produce multiple time-staggereddata sample streams. The communication device also includes a datademultiplexer module adapted to time-deskew and demultiplex the multipletime-staggered data steams. The serial data signal has a multi-gigabitsymbol rate. Each of the time-staggered sampling signals, andcorrespondingly, each of the time-staggered data sample streams, has adata rate below the multi-gigabit symbol rate. The data demultiplexer isadapted to produce a demultiplexed data sample stream representative ofthe serial data signal having the multi-gigabit symbol rate.

In another embodiment, the present invention is directed to a method ina communication device. The method comprises generating a master timingsignal, and generating multiple time-staggered sampling signals based onthe master timing signal. The method further comprises sampling areceived, analog serial data signal in accordance with each of themultiple time-staggered sampling signals, thereby producing multipletime-staggered data sample streams. The method further comprisestime-deskewing the multiple time-staggered data streams to producemultiple time-deskewed data streams, and demultiplexing the multipletime-deskewed data streams.

In yet another embodiment, the present invention is directed to acommunication device on an IC chip. The device is configured to receivemultiple, analog serial data signals. The device comprises a mastertiming generator adapted to generate a master timing signal. The devicealso includes multiple receive-lanes, each configured to receive anassociated one of the multiple serial data signals. Each receive-laneincludes a phase interpolator adapted to produce a sampling signalhaving an interpolated phase, and a data path adapted to sample andquantize the associated serial data signal in accordance with thesampling signal. The device also includes an interpolator control modulecoupled to each receive-lane. The interpolator control module is adaptedto cause the phase interpolator in each receive-lane to rotate theinterpolated phase of the sampling signal in the receive-lane at a ratecorresponding to a frequency offset between the sampling signal and theserial data signal associated with the receive-lane, so as to reduce thefrequency offset between the sampling signal and the serial data signal.

In an even further embodiment, the present invention is directed to amethod in a communication device configured to receive multiple serialdata signals. The method comprises generating a master timing signal,and deriving multiple sampling signals based on the master timingsignal. Each of the multiple sampling signals is associated with one ofthe multiple serial data signals and each of the sampling signals has aninterpolated phase. The method further comprises sampling and quantizingeach of the multiple serial data signals according to the associated oneof the sampling signals. The method also comprises rotating theinterpolated phase of each sampling signal at a rate corresponding to afrequency offset between the sampling signal and the serial data signalassociated with the receive-lane so as to reduce the frequency offsetbetween the sampling signal and the serial data signal. The method alsocomprises rotating each interpolated sampling signal phase independentlyof the other one or more interpolated sampling signal phases.

Terminology

The sampling signal (mentioned above) and the serial data signal areconsidered “phase-aligned” when their respective phases are such thatthe sampling signal causes the serial data signal to be sampled at oracceptably near an optimum sampling time for sampling the serial datasignal.

“Frequency synchronized” or “frequency matched” means the frequencies ofthe sampling signal and serial data signal are related to one anothersuch that the sampling signal and the serial data signal do not tend to“drift” in time relative to one another. For example, once initiallyphase-aligned, the sampling signal and the serial data signal willremain phase-aligned over time as long as the sampling signal and theserial data signal are frequency synchronized. An exemplary frequencymatching condition corresponds to when the frequency of the serial datasignal is an integer multiple (that is, one, two, etc.) of the frequencyof the sampling signal.

When the sampling signal and the serial data signal are “frequencyoffset” from one another, the two signals are not frequencysynchronized. “Nulling” such a frequency offset causes the sampling andserial data signals to be frequency synchronized.

The above defined terms “phase-aligned,” “frequency synchronized,”“frequency matched,” “frequency offset,” and “nulling” shall beconstrued to be consistent with their usage in the followingdescription.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

The present invention will be described with reference to theaccompanying drawings. In the drawings, like reference numbers generallyindicate identical, functionally similar, and/or structurally similarelements. The drawing in which an element first appears is indicated bythe leftmost digit(s) in the reference number.

FIG. 1 is a block diagram of a simple communication system.

FIG. 2 is a block diagram of a portion of an exemplary receiver.

FIG. 3 is a block diagram of a timing recovery module/system.

FIG. 4A is an illustration of an example analog serial data signalwaveform.

FIG. 4B is an illustration of an example symbol of the serial datasignal of FIG. 4A.

FIGS. 4C, 4D, and 4E are illustrations of three different data samplingtime scenarios.

FIG. 5A is a sampled waveform corresponding to the waveform of FIG. 4A.

FIGS. 5B and 5C are example illustrations of data and phase sample timelines.

FIGS. 6A, 6B, and 6C are illustrations of three different sampling timescenarios.

FIG. 7A is a flow diagram of an example method of recovering timinginformation from a serial data signal.

FIG. 7B is a flow chart of an example method expanding on a phaserotating step of the method of FIG. 7A.

FIG. 8 is a block diagram of a phase interpolation environment.

FIG. 9 is a block diagram of a first phase interpolator implementation.

FIG. 10 is a phasor diagram.

FIG. 11 is a schematic of a reference stage circuit that receives abinary control signal.

FIG. 12 is a block diagram of a second phase interpolatorimplementation.

FIG. 13 is a schematic of a reference stage circuit that receives acontrol signal having multiple binary subsignals.

FIGS. 14A and 14B are each schematics of a phase interpolator includinga combining node circuit.

FIG. 15A is a block diagram of a phase interpolator and a phase controlsignal rotator.

FIGS. 15B and 15C are illustrations of alternative implementations of aring of storage cells used in the phase control signal rotator of FIG.15A.

FIGS. 16A-16C are diagrams of phase rings corresponding to the phaseinterpolator and phase control signal rotator of FIG. 15A.

FIG. 17 is an illustration of a frequency offset in a portion of atiming recovery module.

FIG. 18A is an illustration of compensating for the frequency offset ofFIG. 17 in the portion of the timing recovery module.

FIG. 18B is a block diagram of a timing recovery system for frequencysynchronizing a sampling signal to a serial data signal.

FIG. 19 is a block diagram of a phase error processor of FIG. 18B andFIG. 3.

FIG. 20A is a flowchart of a method involving phase and frequencytracking.

FIG. 20B is a flowchart of an example method expanding on a rotatingstep of the method of FIG. 20.

FIG. 20C is a block diagram of an example timing recovery system forsynchronizing sampling and serial data signal frequencies without usinga control signal rotator.

FIG. 20D is a flow chart of an example high level method of controllinga phase interpolator, corresponding to the timing recovery systems ofFIGS. 18B and 20C.

FIG. 20E is a flow chart of an example high level method of frequencysynchronizing a sampling signal to a serial data signal.

FIG. 21 is an illustration of an example of a multiple channelcommunication device constructed on an integrated circuit (IC) chip,according to an embodiment of the present invention.

FIG. 22 is an illustration of a receive-lane corresponding to onechannel of the multiple channel communication device of FIG. 21,according to an embodiment of the present invention.

FIG. 23 is an illustration of various example signal waveforms takenfrom the receive-lane of FIG. 22.

FIG. 24 is a phase circle representing the phases of sampling signals ofFIG. 23.

FIG. 25 is a block diagram of a data demultiplexer module from FIG. 22,according to an embodiment of the present invention.

FIG. 26 is a block diagram of an interpolator control module from FIG.22, according to embodiment of the present invention.

FIG. 27 is a block diagram of a digital data processor of FIGS. 21 and22, according to an embodiment of the present invention.

FIG. 28 is a block diagram of a multiple channel communication device,according to an embodiment of the present invention.

FIG. 29 is a flow chart of an example method of processing a serial datasignal in multiple data paths of a single channel of a serial datareceiver.

FIG. 30 is a flow chart of an example method of frequency synchronizingmultiple data sampling signals to multiple corresponding serial datasignals.

FIG. 31 is an illustration of an example use of atransceiver/communication device of the present invention in an examplesignal router.

FIG. 32 is a block diagram of an alternative phase interpolatorimplementation.

FIG. 33 is a block diagram of another alternative phase interpolatorimplementation.

DETAILED DESCRIPTION OF THE INVENTION I. Overview

FIG. 1 is a block diagram of a simple communication system 100 includinga transmitter 100 and a receiver 102. Transmitter 100 transmits a serialdata signal 104 including, for example, a series of data symbols, toreceiver 102. Serial data signal 104 has a frequency f1 (for example, asymbol baud rate f1) and a phase φ1 both related to a frequency and aphase of an oscillator (not shown) local to transmitter 100.

Receiver 102 samples serial data signal 104 (for example, symbolsincluded in the serial data signal) to recover data from the serial datasignal. Receiver 102 samples serial data signal at sample timesestablished by a sampling signal 106 generated locally at receiver 102.Locally generated sampling signal 106 has a frequency f2 and a phase φ2.

To minimize errors in recovering the data from serial data signal 104,it is desirable that frequencies f2 and f1 match one another, and thatphases φ1 and φ2 are aligned with one another, such that sampling signal106 causes receiver 102 to sample serial data signal 104 at optimumsample times coinciding with occurrences of a maximum Signal-to-Noise(S/N) level of the serial data signal. Often, however, frequency f2 andphase φ2 are respectively offset from frequency f1 and phase φ1 becauseof differences between the respective oscillators used in transmitter100 and receiver 102.

The phase offset between phase φ1 and phase φ2 can cause receiver 102 tosample serial data signal 104 at sub-optimal sample times, while thefrequency offset between frequencies f1 and f2 tends to cause the serialdata signal to “drift” through sampling signal 106. Therefore, suchoffsets can cause errors in recovering the data from serial data signal104. Therefore, it is desirable to compensate for such deleteriousfrequency and phase offsets in receiver 102 in order to optimallyrecover data from serial data signal 104.

II. Exemplary Receiver

FIG. 2 is a block diagram of a portion of an exemplary receiver 200including a timing recovery module 202 of the present invention.Receiver 200 also includes a reference signal generator 204. Timingrecovery module 202 receives serial data signal 104, including, forexample, a series of data symbols. Reference signal generator 204generates a set of reference signals 206 and provides the referencesignal set to timing recovery module 202.

Based on serial data signal 104 and reference signal set 206, timingrecovery module 202 derives a timing/sampling signal 208 used byreceiver 200 to recover data from serial data signal 104.Timing/sampling signal 208 is preferably used as a sampling signal inreceiver 200 to sample symbols included in serial data signal 104.Timing recovery module 202 derives sampling signal 208 such that thesampling signal is phase-aligned with serial data signal 104 and suchthat the frequency of sampling signal 208 matches the frequency (such asa symbol baud rate) of serial data signal 104. In this manner, timingrecovery module 202 recovers timing information (for example, phase andfrequency information) from serial data signal 104 in accordance withthe principles of the present invention, as described in detail below.

FIG. 3 is a block diagram of timing recovery module 202 according to anembodiment of the present invention. Timing recovery module 202 includesa phase controller 302, a phase control signal rotator 304, and a phaseinterpolator 306. Phase controller 302 includes a data path 308, a phasepath 310, a phase detector 312 coupled to the data and phase paths, anda phase error processor 314 coupled to the phase detector.

Also depicted in FIG. 3 is an exemplary block diagram of referencesignal generator 204. Reference signal generator 204 includes areference oscillator 330, a Phase Locked Loop (PLL) 332, and a signalset generator 334. Reference oscillator 330 provides an oscillatorsignal 335 to PLL 332. PLL 332 synthesizes a reference signal 336 basedon oscillator signal 335, and provides the reference signal to signalset generator 334. PLL 332 can include an inductance-capacitance (LC),voltage controlled oscillator, for example. Signal set generator 334generates the set of reference signals 206 based on reference signal336. The reference signals included in reference signal set 206 all havea same frequency but different predetermined phases. Signal setgenerator 334 provides reference signal set 206 to phase interpolator306 of timing recovery module 202. PLL 332 and signal set generator 334can be implemented as a tapped ring oscillator, for example.

Phase interpolator 306 produces timing/sampling signal 208 (mentionedabove in connection with FIG. 2) and a second timing/sampling signal 344offset in phase from sampling signal 208, based on reference signal set206 and a plurality of digital phase control signals 340 applied to thephase interpolator. Sampling signals 208 and 344 each have aninterpolated phase controlled in accordance with digital control signals340. Sampling signal 208 and second sampling signal 344 are alsoreferred to herein as data sampling signal 208 and phase sampling signal344, for reasons that will become apparent from the description below.

Data path 308 includes sampling and quantizing signal processing modulesto sample and quantize serial data signal 104 in accordance with sampletimes established by sampling signal 208. Data path 308 produces a datasignal 346 including sampled and quantized data samples representativeof serial data signal 104. Data path 308 provides data signal 346 tophase detector 312.

Similarly, phase path 310 includes sampling and quantizing signalprocessing modules for sampling and quantizing serial data signal 104 atsample times established by phase sampling signal 344. The samplingtimes established by phase sampling signal 344 are offset in phase fromthe sample times established by data sampling signal 208. Phase path 310produces a second data signal 348 (referred to herein as a phase signal348) including a series of phase samples also indicative of serial datasignal 104. Phase path 310 provides phase signal 348 to phase detector312. Exemplary data and phase paths are described later in connectionwith FIGS. 22 and 28. However, the present invention is not limited tosuch implementations.

Phase detector 312 detects a phase error 350 between data samplingsignal 208 and serial data signal 104 based on the data samples in datasignal 346 and the phase samples in phase signal 348. Phase error 350arises because of an undesirable phase offset between data samplingsignal 208 (and also phase sampling signal 344) and serial data signal104. Thus phase error 350 can be considered a phase error signalindicative of the phase offset between data sampling signal 208 andserial data signal 104.

Phase detector 312 provides phase error 350 to phase error processor314. Phase error processor 314 process phase error 350 to derive one ofa set of rotator control signals or commands 354. Phase error processor314 provides the rotator control command to phase control signal rotator304.

Phase control signal rotator 304 stores the digital phase controlsignals 340 applied to phase interpolator 306, and manipulates the samein response to the rotator control commands 354. Signal rotator 304rotates the plurality of digital phase control signals 340 andcorrespondingly the interpolated phase of data sampling signal 208relative to serial data signal 104 in response to the rotator controlcommands 354, such that the sampling signal and the serial data signalbecome phase aligned with one another.

III. Exemplary Signal Waveforms

FIG. 4A is an illustration of an example waveform 402 of analog serialdata signal 104. Waveform 402 represents a Non-Return-to-Zero (NRZ)signal swinging above (i.e., in a positive direction “+”) and below(i.e., in a negative direction “−”) a zero-line 408, to respectivelyconvey information, such as digital “1s” and “0s.” The serial datasignal represented by waveform 402 includes a series of consecutivesymbols 404 each having a symbol period T. Dashed vertical lines 406 inFIG. 4A represent boundaries between adjacent symbols 404. Receivedsymbols 404 have a “rounded” instead of “squared” appearance because oftransmission band-limiting effects on serial data signal 104.

FIG. 4B is an illustration of an example symbol 410 from waveform 402.An optimum sample time t₀ at which the receiver can sample symbol 410coincides with a maximum amplitude 412 and correspondingly a maximumsymbol S/N of the symbol. In the depicted example, time t₀ coincideswith a mid-point of symbol 410. With reference again to FIG. 3, thepresent invention adjusts the phase of sampling signal 208 to cause datapath 308 to sample each symbol in serial data signal 104 at an optimumdata sample time, such as at sample time t₀ depicted in FIG. 4B.

FIGS. 4C, 4D, and 4E are illustrations of three different data samplingtime scenarios. FIG. 4C is an illustration of an on-time data samplingscenario. With reference to FIG. 3 and FIG. 4C, in the on-time scenario,sampling signal 208 causes data path 308 to sample data symbol 410 at asample time t_(d) (represented as an upward pointing arrow in FIG. 4C)coinciding with optimum sample time t₀, to produce an on-time datasample 414 coinciding with mid-point 412 of symbol 410 (depicted in FIG.4B). Since sample time t_(d) coincides with optimum sample time t₀,sampling signal 208 is considered to be phase-aligned with symbol 410,that is, with serial data signal 104. In other words, there is a minimumacceptable (or preferably a zero-time) offset between sample time t_(d)and optimum sample time t₀.

FIG. 4D is an illustration of an early or leading data samplingscenario. In the leading data sampling scenario, data sample time t_(d)defined by sampling signal 208 precedes (that is, leads) optimum sampletime t₀ because the phase of sampling signal 208 leads the phase ofsymbol 410 in serial data signal 104. In response to this sub-optimalearly phase condition, the present invention retards the phase ofsampling signal 208 relative to serial data signal 104 (and thusreceived symbol 410) so as to align sample time t_(d) with optimumsample time t₀, as depicted in FIG. 4C.

FIG. 4E is an illustration of a late sampling time scenario. In the latesampling scenario, sample time t_(d) follows optimum sample time t₀because the phase of sampling signal 208 lags the phase of serial datasignal 104 (and symbol 410). In response to this sub-optimal laggingphase condition, the present invention advances the phase of samplingsignal 208 so as to align sample time t_(d) with optimum sample time t₀,as depicted in FIG. 4C.

Receiver sampling of serial data signal 104 using data path 308 andphase path 310 in accordance with sampling signals 208 and 344 is nowfurther described by way of example with reference to FIGS. 5A, 5B, and5C. FIG. 5A is a sampled waveform 502 corresponding to serial datasignal waveform 402 of FIG. 4A. Sampled waveform 502 includes a seriesof spaced data samples 504 (depicted as circles superimposed on thewaveform trace) produced by data path 308 in accordance with datasampling signal 208. Sampled waveform 504 also includes a series ofspaced phase samples 506 (depicted as squares superimposed on thewaveform trace) produced by phase path 310 in accordance with phasesampling signal 344.

FIG. 5B is an example data sample timeline 510 established by datasampling signal 208. Data path 308 samples serial data signal 104 atdata sample times, for example, at sample times t_(d1), t_(d2), andt_(d3) (referred to generally as sample time(s) t_(d)), to producecorresponding data samples 504 ₁, 504 ₂, and 504 ₃ included in datasignal 346. Data sampling signal 208 can be a clock wave having anapproximately 50% duty cycle, wherein each sample time t_(d) coincideswith a rising edge of the clock wave, for example. The clockwave canhave higher or lower duty cycles.

FIG. 5C is an example phase sample time line 520 established by phasesampling signal 344. Phase path 310 samples serial data signal 104 atphase sample times of, for example, t_(p1), t_(p2) and t_(p3)(represented as downward pointing arrows and referred to generally asphase sample time(s) t_(p)), to produce corresponding phase samples 506₁, 506 ₂, and 506 ₃ included in phase signal 348. Phase sampling signal344 can be a clock wave having an approximately 50% duty cycle, whereineach sample time t_(p) coincides with a rising edge of the clock wave,for example. The clockwave is not limited to a 50% duty cycle.

When data sampling signal 208 is phase aligned with serial data signal104 as depicted in FIG. 4C, for example, and when the frequency (thatis, pulse repetition interval) of sampling signal 208 matches thefrequency (that is, symbol baud rate) of serial data signal 104,adjacent data sampling times (e.g., t_(d1), t_(d2),) and adjacent phasesampling times (e.g., t_(p1), t_(p2),) are separated by symbol period T.Also, adjacent data and phase sample times (e.g., t_(d1), t_(p1),) areseparated by a half symbol period T/2.

FIGS. 6A, 6B, and 6C are illustrations of sampling time scenarioscorresponding respectively to previously described FIGS. 4C, 4D and 4E,except that phase samples are added to FIGS. 6A, 6B, and 6C. In anon-time sampling scenario illustrated in FIG. 6A, a phase sample 602(for example 506 ₁) precedes data sample 414 (for example 504 ₂), and aphase sample 404 (for example, 506 ₂) follows data sample 414. First andsecond phase samples 602 and 604 coincide with zero-line 408 in theon-time scenario depicted in FIG. 6A. This indicates data sample timet_(d) coincides with optimum sample time t₀, and thus, the phases ofsampling signal 208 and serial data signal 104 are optimally alignedwith one another.

In a leading sampling scenario depicted in FIG. 6B, leading phase sample602 has a negative value while data sample 414 and trailing phase sample604 have positive values. This indicates data sample time t_(d) leadsoptimum sampling time t₀, and thus, the phase of sampling signal 208correspondingly leads the phase of serial data signal 104.

On the other hand, in a lagging sampling time scenario depicted in FIG.6C, leading phase sample 602 and data sample 414 have positive valueswhile trailing phase sample 604 has a negative value. This indicatesdata sample time t_(d) trails optimum sample time t₀, and thus, thephase of sampling signal 208 correspondingly lags the phase of serialdata signal 104.

IV. Exemplary Timing Recovery Method

FIG. 7A is a flow diagram of an example method 700 of recovering timinginformation from an NRZ serial data signal (such as serial data signal104) that can be implemented using timing recovery module 202.

At a first step 702, phase interpolator 306 receives the referencesignals having different phases in reference signal set 206.Interpolator 306 combines in varying relative proportions the referencesignals into data sampling signal 208 in response to the plurality ofdigital phase control signals 340, thereby producing data samplingsignal 208 with a digitally controlled interpolated phase. Phaseinterpolator 306 also produces phase sampling signal 344 in response tothe digital phase control signal such that the phase sampling signal andthe data sampling signal are offset in phase from one another by apredetermined amount corresponding to a fraction (for example, one-half)of a symbol period of serial data signal 104.

At a next step 704, data path 308 samples serial data signal 104 (i.e.,the symbols included in serial data signal 104) at data sample timest_(d) according to the data sampling signal 208 to produce data samplesin data signal 346. Phase path 310 also samples serial data signal 104at phase sample times t_(p) offset in phase relative to the data sampletimes t_(d) to produce phase samples in phase signal 348.

At a next step 706 (depicted in dotted line in FIG. 7A), phase detector312 detects a phase error or phase offset between data sampling signal208 and serial data signal 104 based on the data samples in data signals346 and the phase samples in phase signal 348. Next steps 708, 710 and712 described below collectively represent step 706. At step 708, phasedetector 312 examines the data samples in data signal 346 to detectoccurrences of low-to-high and high-to-low data sample transitions. Suchtransitions occur at symbol boundaries.

At next step 710, phase detector 312 determines whether each of the datasample times t_(d) near to detected data sample transitions (from step706) is early or late with respect to optimum symbol sample time t₀,based on data and phase samples near the detected data sampletransitions.

At next step 712, phase detector 312 derives phase error signal 350indicative of whether each data sample time t_(d) is early or late withrespect to the optimum symbol sample time t₀. Phase detector 312 derivesas the phase error signal an Early decision signal indicating the phaseof sampling signal 208 leads the phase of serial data signal 104 whenthe data sample time t_(d) precedes optimum sample time t₀. Therefore,phase detector 312 produces a series of such Early decision signals overtime while the phase of sampling signal 208 leads the phase of serialdata signal 104.

Alternatively, phase detector 312 derives as the phase error signal 350a Late decision signal indicating the phase of sampling signal 208 lagsthe phase of serial data signal 104 when the data sample time t_(d)follows optimum sample time t₀. Therefore, phase detector 312 produces aseries of such Late decision signals over time while the phase ofsampling signal 208 lags the phase of serial data signal 104.

On the other hand, phase detector 312 tends to produce a series ofrandomly alternating Late and Early decision signals over time whilesampling signal 208 and serial data signal 104 are phase aligned witheach other.

At a next step 720, phase error processor 314 processes the phase errorover time (i.e., over many data and phase samples, and correspondingEarly/Late decision signals) to determine which of the phase controlcommands 354 needs to be asserted. Phase error processor 314 can includean accumulator and/or a filter for accumulating and/or filtering theEarly or Late decision signals included in phase error signal 350, todetermined which of the phase control commands 354 needs to be asserted.The set of phase control commands 354 includes a phase-hold command, aphase-retard command, and a phase-advance command. Phase error processor314 asserts:

a. the phase-hold command when sampling signal 208 and serial datasignal 104 are phase aligned with one another;

b. the phase-retard command when the phase of sampling signal 208 leadsthe phase of serial data signal 104; and

c. the phase-advance command when the phase of sampling signal 208 lagsthe phase of serial data signal 104.

At a next step 730, phase control signal rotator 304 rotates theplurality of digital phase control signals 340, and correspondingly theinterpolated phase of data sampling signal 208, in response to thephase-retard/phase-advance control command asserted by phase errorprocessor 314, such that data sampling signal 208 and serial data signal104 become phase aligned with one another as depicted, for example, inFIG. 6A. Alternatively, signal rotator 304 holds the plurality ofdigital phase control signals 340 in position, thereby preventingrotation of the phase control signals and correspondingly theinterpolated phase of data sampling signal 208, in response to thephase-hold control command when asserted by phase error processor 314.

FIG. 7B is a flow chart of an example method 770 expanding on phaserotating step 730 of method 700. A step 775 is initiated in response tothe phase retard command. At step 775, the interpolated phase ofsampling signal 208 is retarded relative to serial data signal 104.

A step 780 is initiated in response to the phase advance command. Atstep 780, the interpolated phase of sampling signal 208 is advancedrelative to serial data signal 104.

A step 785 is initiated in response to the phase hold command. At step785, the interpolated phase of sampling signal 208 is held at a presentvalue.

V. Phase Interpolator

As described above with reference to FIG. 3 and FIG. 7A, timing recoverymodule 202 includes a phase interpolator 306 that combines referencesignals 206 to generate sampling signals 208 and 344. These samplingsignals are generated by interpolation techniques performed by phaseinterpolator 306. These interpolation techniques can provide samplingsignal phases that span a complete rotation of 360 degrees. These phasesare achieved without the use of conventional techniques, such astime-delays.

In addition to the exemplary timing recovery and receiver applicationsdescribed herein, the phase interpolation techniques of the presentinvention may be used in other applications.

An exemplary phase interpolator environment is now described. FIG. 8 isa block diagram of a phase interpolation environment 800. Environment800 includes a phase interpolator 801, such as phase interpolator 306,and a stage controller 806, such as phase control signal rotator 304 orother controller. The controller 806 is not limited to a control signalrotator.

Interpolator 801 includes a plurality of reference stages 802 a-d thatare each coupled to stage controller 806, and a combining node 804 thatis coupled to each of reference stages 802. As shown in FIG. 8, eachreference stage 802 receives a corresponding reference signal 820. Thesereference signals are each periodic waveforms that each have a distinctphase. Examples of periodic waveforms include sinusoid, rectangularwaveforms, trapezoidal waveforms, and other similar periodic signals.

In addition, each reference stage 802 receives a corresponding controlsignal 822 from stage controller 806. As shown in FIG. 8, referencestage 802 a receives control signal 822 a, reference stage 802 breceives control signal 822 b, and reference stage 802 c receivescontrol signal 822 c.

Each reference stage 802 generates a component signal 824 from itscorresponding reference signal 822 according to a scaling factor that isthe ratio of a component signal 824 magnitude to its correspondingreference signal 820 magnitude. A reference stage 802 scaling factor isdetermined by its corresponding control signal 822. For example,reference stage 802 a generates component signal 824 a from referencesignal 822 a according to a scaling factor determined by control signal822 a.

These scaling factors control the magnitude of corresponding componentsignals 824. This controlled magnitude may be zero. Thus, controlsignals 822 may scale as well as activate and deactivate correspondingcomponent signals 824.

Component signals 824 are each sent to combining node 804. As shown inFIG. 8, reference stage 802 a generates a component signal 824 a,reference stage 802 b generates a component signal 824 b, referencestage 802 c generates a component signal 824 c, and so on.

Combining node 804 combines each of component signals 824 to produce anoutput signal 826. This combining includes summing each of theindividual component signals 824 (some of which may have a magnitudeequal to zero). As a result of this combining, output signal 826 is aperiodic waveform having a phase that is derived from the phases ofcomponent signals 824. This derivation is referred to herein as phaseinterpolation.

Stage controller 806 generates stage control signals 822 in response toan interpolation command 828 that is received from a master systemcontroller (not shown), such as rotator control commands 354 receivedfrom phase error processor 314. Exemplary details regardinginterpolation command 828 are provided in greater detail below.

As described above, each reference stage 802 generates a componentsignal 824 from a reference signal 820 having a distinct phase. Thesegenerated component signals 824 each have a distinct phase that isdetermined by the corresponding reference signal 820 phase. For example,a component signal 824 may have the same or substantially the same phaseas its corresponding reference signal 820. Alternatively, a componentsignal 824 may have a phase that is offset by a predetermined phaseshift from the corresponding reference signal 820 phase.

Through phase interpolation, the present invention can provide acomplete range of phases (i.e., 360 degrees) for output signal 826. Thiscomplete range is provided through the deployment of more than tworeference stages 802 and a strategic predetermined selection ofreference signal 820 phases.

FIG. 9 is a block diagram of a first phase interpolator 801implementation. This implementation includes four reference stages 802a-d that receive reference signals 820 a-d, respectively. Each ofreference signals 820 a-d has a distinct, predetermined phase. As shownin FIG. 9 by way of example, reference signal 820 a has a phase 910 a ofzero degrees, reference signal 820 b has a phase 910 b of 90 degrees,reference signal 820 c has a phase 910 c of 180 degrees, and referencesignal 820 d has a phase 910 d of 270 degrees. Thus, the implementationof FIG. 9 includes four reference signals 820 having phases 910 that areseparated at intervals of 90 degrees.

In addition, FIG. 9 illustrates that each reference stage 802 includes ascaling module 902, and a conversion module 904 that is coupled toscaling module 902. Each conversion module 904 receives and converts areference signal 820 into a corresponding component signal 824 accordingto a scaling factor. Scaling module 902 establishes this scaling factorin response to its corresponding control signal 822. Details regardingimplementations of scaling module 902 and conversion module 904 areprovided below.

The phase interpolator 801 implementation shown in FIG. 9 receivesbinary control signals 822 that are capable of having two distinctvalues (i.e., 0 and 1). Accordingly, FIG. 9 shows phase interpolator 801having an operational state where control signals 822 a, 822 c, and 822d have values of 0, and control signal 822 d has a value of 1.

FIG. 10 is a phasor diagram that illustrates the phase interpolationcapabilities of the phase interpolator 801 implementation shown in FIG.9. This implementation is capable of generating output signal 826 havingone of eight possible phases. These eight possible phases are spaced atintervals of 45 degrees, and span a complete rotation of 360 degrees.Phasor diagram 1000 includes phasors 1002, 1006, 1010, and 1014. Thesephasors have the same phases as reference signal phases 910 a, 910 b,910 c, and 910 d, respectively.

In addition, phasor diagram 1000 includes phasors 1004, 1008, 1012, and1016. These phasors have phases that are between reference phases 910a-d. As shown in FIG. 10, phasor 1004 has a phase of 45 degrees, phasor1008 has a phase of 135 degrees, phasor 1012 has a phase of 225 degrees,and phasor 1016 has a phase of 315 degrees. These “between” phases areestablished through combining two component signals 824 at combiningnode 804.

Table 1, below, shows how the values of control signals 822 a through822 d determine which of the phasors in FIG. 3 represents output signal826.

TABLE 1 Control Signals Output Signals 2922a 2922b 2922c 2922d PhasorPhase 0 0 0 1 1002 0 0 0 1 1 1004 45 0 0 1 0 1006 90 0 1 1 0 1008 135 01 0 0 1010 180 1 1 0 0 1012 225 1 0 0 0 1014 270 1 0 0 1 1016 315

Thus, the phase interpolator 801 implementation of FIG. 9 can adjust thephase of output signal 826 among eight distinct phases.

FIG. 11 is an exemplary schematic of a reference stage 802 circuit thatreceives a binary control signal 822. Thus, this circuit may be employedin the phase interpolator 801 implementation of FIG. 9. In this circuit,reference signals 820 and component signals 824 are each differentialsignal pairs that have an in-phase signal and a 180 degrees out-of-phasesignal. Except for a 180 degrees phase shift, these signals are thesame. As shown in FIG. 11, reference signal 820 includes an in-phasesignal 1120 and an out-of-phase signal 1122. Similarly, component signal824 includes an in-phase signal 1124 and an out-of-phase signal 1126.These signals are time varying voltage signals.

As described above with reference to FIG. 9, reference stage 802includes a scaling module 902 and a conversion module 904. As shown inFIG. 11, conversion module 904 includes two N channel metal oxidesemiconductor (NMOS) field effect transistors (FETs) 1102 and 1104 thateach have drain, source, and gate terminals. However, conversion module904 may employ other transconductance devices.

Scaling module 902 includes a current digital to analog converter (IDAC)1106 that is coupled to the source terminals of FETs 1102 and 1104.

Scaling module 902 receives binary control signal 822. When binarycontrol signal 822 has a value of 1, IDAC 1106 operates as a currentgenerator that enables a current 1128 to flow through the drain andsource terminals of FETs 1102 and 1104. However, when binary controlsignal 822 has a value of 0, IDAC 1106 does not enable current 1128 toflow (i.e., current 1128 has zero magnitude).

The flow of current 1128 enables reference signal 820 to be convertedinto corresponding component signal 824. That is, source current 1128enables the conversion of differential reference signals 1120 and 1122into differential component signals 1124 and 1126, respectively. Thisconversion is performed according to a specific scaling factor.

Differential component signals 1124 and 1126 are electrical currentsignals that are combined at combining node 804 with differentialcomponent signals from other reference stages 802. This combininggenerates output signal 826. An exemplary combining node 804 circuitschematic is described below with reference to FIG. 14.

As described above, the phase interpolator 801 implementation of FIG. 9is capable of producing eight different phases for control signal 826 ata granularity of 45 degrees. However, the present invention may achievefiner phase granularity through implementations where each controlsignal 822 is capable of having more than two distinct values.

FIG. 12 is a block diagram showing an implementation of phaseinterpolator 801 that receives control signals 822 capable of havingmore than two distinct values. This implementation enables output signal826 to have a greater number of phases than the implementation of FIG.9. The FIG. 12 implementation of phase interpolator 801 includes aplurality of reference stages 802′.

Unlike reference stages 802 of FIG. 9, each reference stage 802′receives a composite control signal 822′ that includes a plurality ofbinary subsignals 1220. For example, reference stage 802 a′ receivescomposite control signal 822 a′, which includes subsignals 1220 a-d.These subsignals 1220 each contribute to the value of the correspondingcomposite control signal 822′. For example, subsignals 1220 a-dcontribute to the value of composite control signal 822 a′.

For purposes of convenience, only reference stage 802 a′ will bedescribed in detail. However, the other reference stages 802′ shown inFIG. 12 may include identical or similar features. Reference stage 802a′ includes a plurality of scaling modules 902. In particular, theimplementation of FIG. 12 shows four scaling modules 902 a-902 d.However, any number may be employed. In addition, reference stage 802 a′includes a conversion module 904 that is coupled to each of scalingmodules 902 a-d.

Scaling modules 902 a-d each receive a respective one of subsignals 1220a-d. As shown in FIG. 12, scaling module 902 a receives subsignal 1220a, scaling module 902 b receives subsignal 1220 b, scaling module 902 creceives subsignal 1220 c, and scaling module 902 d receives subsignal1220 d.

Each of scaling modules 902 a-d provide an individual contribution tothe reference stage 802′ scaling factor. These individual contributionsare based on the value of the corresponding control subsignal 1220. Asdescribed above, scaling factor is the ratio of a component signal 824magnitude to its corresponding reference signal 820 magnitude.Accordingly, the aggregate sum of control signals 1220 a-d (alsoreferred to herein as the value of composite control signal 822′)determines the reference stage 802 a′ scaling factor according to apredetermined relationship. According to one such relationship, thereference stage 802 a′ scaling factor increases with the value ofcomposite control signal 822′.

Since subsignals 1220 a-d are each binary signals, aggregate controlsignals 822′ can have five distinct values. Thus, reference stage 802 a′can generate component signal 824 a from reference signal 820 aaccording to five different scaling factors. One of these scalingfactors may be equal to zero, thereby causing corresponding componentsignal 824 a to also have a magnitude of zero. Thus, the phaseinterpolator 801 implementation of FIG. 11 generates component signals824 a-d that each may have one of five different magnitudes. These fivedifferent magnitudes advantageously provide a number of attainablecontrol signal 826 phases across a 360 degrees range that is greaterthan the eight phases achievable with the phase interpolator 801implementation of FIG. 9.

FIG. 13 is a schematic of a reference stage 802′ circuit that receives acontrol signal having multiple binary subsignals. Thus, this circuit maybe employed in the phase interpolator 801 implementation of FIG. 12,which receives a plurality of control subsignals 1220 a-d. In thiscircuit, reference signals 820 and component signals 824 are eachdifferential signal pairs that have an in-phase signal and a 180 degreesout-of-phase signal. Except for a 180 degrees phase shift, these signalsare the same. As shown in FIG. 13, reference signal 820 includes anin-phase signal 1320 and an out-of-phase signal 1322. Similarly,component signal 824 includes an in-phase signal 1324 and anout-of-phase signal 1326. These signals are time varying voltagesignals.

As described above with reference to FIG. 12, reference stage 802includes a plurality of scaling modules 902 a-d and a conversion module904. Conversion modules 904 includes two N channel metal oxidesemiconductor (NMOS) field effect transistors (FETs) 1302 and 1304 thateach have drain, source, and gate terminals. However, conversion modules904 may employ other transconductance devices.

Scaling modules 902 a-d each include an IDAC 1306, shown in FIG. 13 asIDACs 1306 a-d. IDACs 1306 a-d are each coupled to the source terminalsof FETs 1302 and 1304.

Each of IDACs 1306 a-d receives a respective one of binary controlsubsignals 1220 a-d and, enables a corresponding current 1328 to flowfrom the source terminals of FETs 1302 and 1304 when the respectivecontrol subsignal 1220 has a value of 1. For example, IDAC 1306 aenables a current 1328 a to flow when subsignal 1220 a equals 1.However, when a control subsignal 1220 has a value of 0, thecorresponding IDAC 1306 does not enable corresponding current 1328 toflow (i.e., corresponding current 1328 has zero magnitude).

Currents 1328 a-d each contribute to an aggregate current 1330. Thevalue of aggregate current 1330 depends on the number of IDACs 1306 thatare receiving a subsignal 1220 having a value of 1. As aggregate current1330 increases, so does the scaling factor associated with theconversion of differential reference signals 1320 and 1322 intodifferential component signals 1324 and 1326, respectively.

Component signals 1324 and 1326 are electrical current signals. Thesecurrent signals are combined at combining node 804 with componentsignals from other reference stages 802. This combining generates outputsignal 826. An exemplary combining node 804 circuit schematic isdescribed below with reference to FIGS. 14A and 14B.

FIGS. 14A and 14B are schematics illustrating a combining node 804circuit coupled to various reference stage 802 implementations. FIG.14A, illustrates this combining node 804 coupled to the four referencestage 802 circuit of FIG. 11. However, FIG. 14B illustrates thiscombining node 804 circuit coupled to four reference stage 802′ circuitsof FIG. 13.

The combining node 804 circuit of FIGS. 14A and 14B includes a firstresistor 1402, and a second resistor 1404. Resistors 1402 and 1404 areeach coupled to a voltage node 1406, such as a Vdd rail. In addition,resistors 1402 and 1404 are coupled to reference stages 820 a-d. Asshown in FIGS. 14A and 14B, resistor 1402 is coupled to a FET 1102within each reference stage 820. Similarly, resistor 1404 is coupled toa FET 1104 within each reference stage 120.

The combining node 804 circuit of FIGS. 14A and 14B also includes afirst output node 1408 and a second output node 1410. Output nodes 1408and 1410 provide output signal 826 in the form of a differential signalhaving an in-phase output signal 1420 and an out-of-phase output signal1422. Output signals 1420 and 1422 are voltage signals measured inrelation to a reference voltage, such as ground.

As described above with reference to FIGS. 11 and 13, each referencestage 802 includes a conversion module 904 that can generate acorresponding component signal 824 in the form of electrical currentsignals. Examples of such electrical current signals include signals1124 and 1126, and signals 1324 and 1326.

In the FIG. 14A combining node 804 circuit, current signals 1124 a-d and1126 a-d contribute to a voltage drop across resistors 1402 and 1404,respectively. Similarly, in the FIG. 14B combining node 804 circuit,current signals 1324 a-d and 1326 a-d contribute to a voltage dropacross resistors 1402 and 1404, respectively. In FIGS. 14A and 14B,output signal 826 (i.e., output signals 1420 and 1422), is based onthese voltage drops.

VI. Phase Rotation

FIG. 15A is a block diagram of phase interpolator 306 and phase controlsignal rotator 304 according to an embodiment of the present invention.For exemplary purposes, the embodiment of phase interpolator 306depicted in FIG. 15A corresponds to the phase interpolator described inconnection with FIGS. 12 and 14B. Also, the embodiment of control signalrotator 304 depicted in FIG. 15A is compatible with the depicted phaseinterpolator embodiment. Other embodiments of phase interpolator 306(and correspondingly, of control signal rotator 304) are possible, aswould be apparent to one of ordinary skill in the relevant art(s) afterreading the description provided herein. For example, phase interpolator306 can be implemented in accordance with the phase interpolatorembodiments described above in connection with FIGS. 9, 11, and 14A, andbelow in connection with FIGS. 32 and 33.

Phase control signal rotator 304 (also referred to as signal rotator304) receives phase control command set 354 from phase error processor314. As mentioned above, and as depicted in FIG. 15A, phase controlcommand set 354 includes a phase-advance command 354 a, a phase-retardcommand 354 b, and a phase-hold command 354 c. Phase-advance command 354a can be considered as a rotate-left command (that is, as a command torotate the phase of sampling signal 208 in a counter-clockwise directionto advance its phase). Phase-retard command 354 a can be considered as arotate-right command (that is, as a command to rotate the phase ofsampling signal 208 in a clockwise direction to retard its phase).

Signal rotator 304 manipulates the digital phase control signals 340 inaccordance with an asserted one of phase control commands 354, andprovides the so manipulated digital phase control signals 340 to phaseinterpolator 306, as will be described in further detail below. Signalrotator 304 includes a plurality of storage cells 1502 arranged in aring configuration, generally referred to as a ring of storage cells1504. The ring of storage cells 1504 includes a plurality of ringsegments 1506 a, 1506 b, 1506 c, and 1506 d connected to one another bysignal lines 1508 a-1508 d in the ring configuration, as depicted inFIG. 15A. Each of the ring segments includes a plurality of theindividual storage cells 1502. Each of the storage cells 1502 stores acorresponding one of the plurality of digital phase control signals 340.In one arrangement, the ring of storage cells 1504 is implemented as acircular shift register responsive to a shift-left, a shift-right, and ashift-enable control input (corresponding to commands 354 a, 354 b, and354 c, for example).

Each one of the digital phase control signals 340 can be a digital(i.e., logical) “1” or a digital “0,” for example. Therefore, each ofthe storage cells 1504 can store a digital “1” or a digital “0,”representing one of the digital phase control signals at any given time.An exemplary arrangement of digital phase control signals stored in ring1504 is depicted in FIG. 15A, wherein each of the storage cells 1502included in ring segment 1506 a is a logical “1,” while the remainder ofthe digital phase control phase signals stored in the storage cells ofthe other ring segments 1506 b-1506 d are all logical “0s.”

In the arrangement described above, digital phase control signals 340are divided among a plurality of digital phase control signal sets 340a, 340 b, 340 c, and 340 d. Each of the signal sets 340 a-340 dcorresponds to a respective one of ring segments 1506 a, 1506 b, 1506 c,and 1506 d. In other words, the storage cells included in ring segment1506 a collectively provide digital phase control signal set 340 a tophase interpolator 306, and so on.

Phase interpolation is described above in connection with FIGS. 8-14,and is now described briefly again for purposes of convenience. Phaseinterpolator 306 is capable of bringing about phase shifts havinggranularity that is finer than 45 degrees. Thus, phase interpolator 306includes reference stages 802 a′, 802 b′, 802 c′, and 802 d′, asdescribed above with reference to FIGS. 12, 13, and 14B.

Each of the reference stages 802 a′-802 d′ receives a corresponding oneof the set of digital phase control signals 340 a-340 d (for example,ring segment 1506 a of ring 1504 provides digital phase control signalset 340 a to reference stage 802 a′, and so on). These phase controlsignal sets are discrete signals capable of having more than twodistinct values. Control signals sets 340 correspond to control signals822′ in FIG. 12.

Phase interpolator 306 also receives reference signal set 206 (820 inFIG. 12) from reference signal generator 216 (see FIG. 3). Referencesignal set 206 includes reference signals 206 a, 206 b, 206 c, and 206d. Reference signals 206 a, 206 b, 206 c and 206 d each have respectiverelative reference phases of 0°, 90°, 180°, and 270°, for example.Reference stages 802 a′-80 d′ respectively derive component signals 824a-824 d, each having a phase based on (for example, equal to) acorresponding one of the reference signals 206 a-206 d. For example,each of reference stages 802 a′-802 d scales an amplitude of acorresponding one of reference signals 206 a-206 d in response to thecorresponding one of signal sets 340 a-340 d, to produce a correspondingone of the component signals 824 a-824 d, in the manner describedpreviously. Combining node 804 combines the component signals 802 a′-802d′ (representing scaled versions of respective reference signals 206a-206 d) into output signal 826, which is sampling signal 208 in thiscontext.

Therefore, phase interpolator 306 can be considered as combining thesignals in reference signal set 206, having the different phases, intosampling signal 208 having the interpolated phase. Phase interpolator306 varies the relative proportions of the reference signals so combinedin response to the plurality of digital phase control signal 340 appliedto the interpolator. More specifically, each of signal sets 340 a-340 dcontrols the relative proportion of the corresponding one of thereference signals 206 a-206 d combined into sampling signal 208 byinterpolator 306. It is to be understood that “relative proportion”refers to a proportion value ranging between a minimum value (such aszero, whereby a reference signal does not contribute to the interpolatedphase) and a maximum value.

When phase error processor 314 asserts rotate-left command 354 a (thatis, the phase-advance command), signal rotator ring 1504 concurrentlyshifts-left (that is, in the direction indicated by an arrow L) each oneof the digital phase control signals 340 from a present storage elementto an adjacent next storage element to the left of the present storageelement, in response to the command. Therefore, ring 1504 rotates all ofthe digital phase control signals 340 in counter-clockwise direction L.In response, phase interpolator 306 correspondingly rotates theinterpolated phase of sampling signal 208 in the counter-clockwisedirection (in a direction of decreasing phase), thereby advancing thephase of sampling signal 208 relative to serial data signal 104.

When phase error processor 314 asserts rotate-right command 354 b (thatis, the phase-retard command), ring 1504 concurrently shifts-right (thatis, in a clockwise direction indicated by an arrow R) each one of thedigital phase control signals 340 from the present storage element to anadjacent next storage element to the right of the present storageelement, in response to the command. Therefore, ring 1504 rotates all ofthe digital phase control signals 340 in clockwise direction R. Inresponse, phase interpolator 306 correspondingly rotates theinterpolated phase of sampling signal 208 in the clockwise direction (ina direction of increasing phase), thereby retarding the phase ofsampling signal 208 relative to serial data signal 104.

Phase-hold command 354 c overrides either of commands 354 a and 354 b.Therefore, when phase error processor 314 asserts phase-hold command 354c, ring 1504 holds all of the digital control signals in each presentstorage element, in response to the command. In other words, phase-holdcommand 354 c prevents all of the digital phase control signals andcorrespondingly the interpolated phase of sampling signal 208 fromrotating.

Phase-advance and -retard commands 354 a and 354 b can be implemented aspulsed commands. As such, a single, pulsed phase-advance command 354 a(also referred to as a phase-advance pulse 354 a) causes an incrementalshift-left of one position, and correspondingly, an incremental phaseadvance, as described above. Similarly, a single, pulsed phase-retardcommand 354 b causes an incremental shift-right of one position, andcorrespondingly, an incremental phase retardation, as is also describedabove. Thus, the interpolated phase of sampling signal 208 can beincrementally rotated clockwise or counter-clockwise through a range of360° by successively pulsing phase-retard and phase-advance commands 354b and 354 a, respectively. The rate at which the interpolated phase ofsampling signal 208 rotates corresponds to the repetition rate of pulsedphase-retard and phase-advance commands 354 b and 354 a.

FIGS. 15B and 15C, described below, are illustrations of alternativeimplementations of ring 1504. FIG. 15B is a block diagram of ring 1504implemented as a circular shift register 1550. Shift register 1550includes linearly arranged storage cells 1502 linked together tocollectively form the ring configuration. Shift register 1550 includesleft and right end cells (not labeled), and a signal line 1530 couplingthe end cells together.

FIG. 15C is a block diagram of ring 1504 implemented as an array ofstorage cells 1560. Array 1560 includes storage cells 1502 arranged as amatrix of rows and columns, as depicted in FIG. 15C. Alternativeimplementations of ring 1504 are possible, as would be apparent to oneof ordinary skill in the relevant art after reading the descriptionprovided herein.

VII. Phasor Diagrams

FIG. 16A is an illustration of an exemplary phase ring 1600 useful fordescribing phase rotation in the present invention. Phase ring 1600includes phase segments 1606 a, 1606 b, 1606 c, and 1606 d correspondingto ring segments 1506 a-1506 d of ring 1504, and to interpolator stages802 a′-802 b′, depicted in FIG. 15A. Each of the phase segments 1606a-1606 d is divided into individual, contiguous phase cells 1608, eachrepresentative of a discrete phase value. The phase cells 1608 of eachof phase segments 1606 a, 1606 b, 1606 c, and 1606 d straddle respectivephase values of 0°, 90°, 180°, and 270°. (corresponding to the phases ofreference signals 206 a-206 d) superimposed around phase ring 1600. Thedistribution of digital phase control signals (logical “1s” and “0s”)depicted within phase cells 1608 corresponds to the exemplarydistribution of the same control signals stored in ring 1504 of signalrotator 304, depicted in FIG. 15A.

The distribution of digital phase control signals within phase cells1608 illustrated in FIG. 16A indicates the relative proportion of thereference phases 0°, 90°, 180°, and 270° included in a resultant phasor1620 representing the resultant phase of interpolated sampling signal208. As depicted in FIG. 16A, a set of four contiguous logical “1s” 1610resides in phase segment 1606 a, while logical “0s” reside elsewhere.Therefore, reference or component phase 0° is turned full-on, while allof the other phases are turned-off. That is, the relative proportions ofthe reference phases are such that phase 0° is at a maximum value insampling signal 208, while the other phases are at minimum values (ofzero, for example). Therefore, the phase of sampling signal 208 outputby interpolator 306 is 0°.

FIG. 16B is an illustration of phase ring 1600 after signal rotator 304shifts phase control signals 340 from the positions depicted in FIG. 15A(and correspondingly, in FIG. 16A) two positions to the right (that is,clockwise) in response to two consecutive phase-retard pulses (i.e.,commands) 354 b. The consecutive phase-retard pulses 354 b arerepresented as consecutive clockwise pointing arrows 354 b in FIG. 16B.In accordance with the distribution of control signals 340 depicted inFIG. 16B, each of component phases 0° and 90° is at half of its maximumvalue (since the four logical “1s” 1610 are distributed such that twoare within phase segment 1606 a corresponding to phase 0° while theother two are within phase segment 1606 b corresponding to phase 90°),while all other phases are turned off. Therefore, interpolator 314produces sampling signal 208 with an interpolated phase 1620 of 45°(half-way between 0° and 90°).

FIG. 16C is an illustration of phase ring 1600 after signal rotator 304shifts phase control signals 340 from the positions depicted in FIG. 15A(and correspondingly, in FIG. 16A) two positions to the left (that is,counter-clockwise) in response to two consecutive phase-advance pulses(i.e., commands) 354 a. The consecutive phase-advance pulses 354 a arerepresented as consecutive counter-clockwise pointing arrows 354 a inFIG. 16C. In accordance with the distribution of control signals 340depicted in FIG. 16C, each of component phases 0° and 270° is at half ofits maximum value (since the four logical “1s” 1610 are distributed suchthat two are within phase segment 1606 a while the other two are withinphase segment 1606 d), while all other phases are turned off. Therefore,interpolator 314 produces sampling signal 208 with an interpolated phase1620 of 315° (half-way between 0° and 360°).

In the exemplary configurations depicted in FIGS. 15A and 16A-16C, phaseinterpolator 314 can produce sixteen different phases ranging from 0° to270° with a phase resolution of approximately 22° (360°/16≈22°).

The density of phase control signal logical “1s” within the phase ringremains constant as the digital phase control signals 340 andcorrespondingly the interpolated phase is rotated. As a result, samplingsignal 208 advantageously maintains a constant amplitude as the phase ofthe sampling signal varies over a range of 360°. For example, withreference to the exemplary circuits shown in FIGS. 13 and 14B, aconstant density of logical “is” maintains a constant number of currents1328. This, in turn, provides constant amplitude output signals 826.

VIII. Frequency Synchronization

Interpolator 306 produces sampling signal 208 at a sampling frequencyω_(s) (where angular frequency ω_(s)=2πf_(s)) based on a frequency ω_(r)common to each reference signal in the set of reference signals 206 fromreference signal generator 304 (that is, each of the reference signalshas the reference frequency ω_(r)). In the embodiment of interpolator306 described above in connection with FIG. 15A, frequency ω_(s) ofinterpolated sampling signal 208 is equal to reference frequency ω_(r)of each of the reference signals in reference signal set 206 (that is,ω_(s)=ω_(r)).

FIG. 17 is an illustration of a portion of timing recovery module 202corresponding to when an undesirable angular frequency offset Δω existsbetween serial data signal 104 and sampling signal 208. As depicted inFIG. 17, serial data signal 104 has an angular frequency ω_(d)established by a transmit oscillator (not shown) remote from andindependent of reference signal generator 304 in the present invention.Because of differences between the remote transmit oscillator andreference oscillator 304, frequency ω_(d) and reference frequency ω_(r)may be offset from one another by frequency offset Δω (for example,ω_(d)=ω_(r)+Δω), as depicted in FIG. 17. Therefore, serial datafrequency ω_(d) and sampling frequency ω_(s) are correspondingly offsetfrom one another by the same frequency offset, Δω.

As mentioned above, it is desirable for sampling frequency ω_(s) tomatch serial data frequency ω_(d) (for example, such that ω_(d)=n·ω_(s),where n is an integer greater than zero), whereby once serial datasignal 104 and sampling signal 208 are phase aligned with each other,they remain phase aligned over time. Therefore, timing recovery module202 of the present invention adjusts sampling frequency ω_(s) tocompensate for the above mentioned frequency offset Δω, to thereby matchthe frequency of sampling signal 208 to that of serial data signal 104.The present invention adjusts sampling frequency ω_(s) in the mannerdescribed below.

Interpolated sampling signal 208 has a frequency ω_(s) (based onreference frequency ω_(r)) and an interpolated phase φ_(I) (for example,see phasor 1620 in FIGS. 16A-16C). While the interpolated phase φ_(I) ofsampling signal 208 is maintained at or dithered around aconstant/average phase value, sampling signal frequency ω_(s) iscorrespondingly maintained at a base frequency equal to referencefrequency ω_(r). However, since frequency is the derivative of phase(that is, ω=dφ/dt, where φ is phase), interpolator 306 can repetitivelyrotate interpolated phase φ_(I) through 360° at a predetermined rate tofrequency shift sampling frequency ω_(s) away from the base frequencyω_(r). The magnitude of the frequency shift, Δω_(I), is governed by theequation:Δω_(I) =dφ _(I) /dt,where dφ_(I)/dt represents the rate at which phase is rotated.

Accordingly, the sampling frequency ω_(s) of sampling signal 208 isgoverned by the equation:ω_(s)=ω_(r) ±dφ _(I) /dt, or equivalentlyω_(s)=ω_(r)±Δω_(I).

Therefore, the present invention can rotate phase φ_(I) of samplingsignal 208 at different rates to correspondingly produce differentsampling frequencies ω_(s).

FIG. 18A is an illustration of a portion of timing recovery module 202,wherein phase rotation as described above is used to compensate for afrequency difference between serial data signal 104 and referencesignals 206 (that is, phase rotation is used to match the frequency ofsampling signal 208 to that of serial data signal 104). Serial datasignal 104 has a frequency ω_(d)=ω_(r)+Δω. Timing recovery module 202causes interpolator 306 to rotate interpolated phase φ_(I)counter-clockwise in the direction indicated by an arrow 1806 at a ratecorresponding to Δω, such that dφ_(I)/dt=Δω₁=Δω. Therefore, interpolator306 produces sampling signal 208 at frequency ω_(s)=ω_(r)+dφ_(I)/dt, orequivalently ω_(s)=ω_(r)+Δω, such that sampling signal 208 and serialdata signal 104 have matching frequencies.

FIG. 18B is a block diagram of a timing recover system 1810 forfrequency synchronizing sampling/timing signal 208 with serial datasignal 104, according to an embodiment of the present invention. FIG.18B is similar to FIG. 3. Timing recover system 1810 includes phaseinterpolator 306 coupled to a controller 1820 for controlling the phaseinterpolator. Controller 1820 includes data and phase paths 308 and 310,phase detector 312, phase error processor 314, and phase control signalrotator 304. Controller 1820 applies control signals 340 to phaseinterpolator 306 to control the interpolated phase of sampling signal208 (and 344). Controller 1820 includes phase error processor 314 toderive an estimate of a frequency effort between sampling signal 208 andserial data signal 104, as will be described in further detail below.Controller 1820 manipulates control signals 340 in response to thefrequency offset, to cause phase interpolator 306 to rotate theinterpolated phase of sampling signal 208 at a rate corresponding thefrequency offset, so as to reduce the frequency offset between serialdata signal 104 and sampling signal 208.

FIG. 19 is a block diagram of phase error processor 314 according to anembodiment of the present invention. Phase error processor 314 includesa short-term phase error processor 1904, a frequency offset estimator1906 (also referred to as a long-term phase processor 1906), and arotate command generator 1908. Short-term processor 1904 and frequencyoffset estimator 1906 receive phase error 350 from phase detector 312.

Short-term processor 1904 integrates phase errors over a relativelyshort time period, and thus responds relatively rapidly to changes inphase between sampling signal 208 and serial data signal 104. Processor1904 derives a phase adjust signal 1910 in response to theaforementioned short-term phase changes. Processor 1904 provides thephase adjust signal 1910 to rotate command generator 1908.

On the other hand, frequency estimator 1906 integrates phase errors overa relatively long period of time (for example, in comparison toshort-term processor 1904), and thus, responds relatively slowly tochanges in phase between sampling signal 208 and serial data signal 104.Frequency estimator 1906 examines changes in phase error signal 350 overtime t₀ derive an estimate of a frequency offset, for example, Δω,between serial data signal 104 and sampling signal 208 (which may resultfrom a corresponding frequency offset between serial data signal 104 andreference signals 206). Frequency estimator 1906 provides a signal 1912indicative of frequency offset estimate Δω to rotate command generator1908.

In alternative arrangements, the functions performed by frequencyestimator 1906 and short-term processor 1904 can be combined into asingle logic block. Alternatively, frequency estimator 1906 canintegrate signal 1910 output by short-term processor 1904, to producesignal 1912. Also, short-term processor 1904 and frequency estimator1906 can be implemented as accumulators, such that signals 1910 and 1912include accumulator over- and under-flow conditions. Other embodimentsof phase error processor 314 are possible as would be apparent to one ofordinary skill in the relevant art(s), after reading the descriptionprovided herein.

Rotate command generator 1908 derives rotate commands 354 (describedabove) based on phase adjust signal 1910 and frequency offset estimatesignal 1912. Rotate command generator 1908 can be part of one or both ofblocks 1904 and 1906. In one embodiment, rotate command generator 1908generates pulsed phase-advance and phase-retard commands 354 a and 354 b(described above) in response to signals 1910 and 1912. In such anembodiment, rotate command generator 1908 generates pulsed commands 354at a repetition rate based on the frequency offset estimate Δω providedin signal 1912. This causes digital control signals 340 andcorrespondingly the phase of sampling signal 208 to rotate at a ratebased on (for example, equal to) the frequency offset Δω. On the otherhand, phase adjust signal 1910 tends to perturbate the above mentionedrepetition rate and correspondingly the phase rotation rate of samplingsignal 208, in response to short-term phase errors. In the abovedescribed manner, timing recovery module 202 can adjust sampling signalfrequency ω_(s) to match serial data frequency ω_(d).

Other embodiments of rotate command generator 1908 are possible as wouldbe apparent to one of ordinary skill in the relevant art(s), afterreading the description provided herein.

Timing recovery module 202 implements a phase and frequency locked (thatis, tracking) loop, including phase controller 302, phase control signalrotator 304, and phase interpolator 306, all described previously. Thephase and frequency locked loop causes the sampling signal phase andfrequency to track the serial data signal phase and frequency, wherebysampling signal 208 and serial data signal 104 remain phase-aligned andfrequency synchronized over time.

Short-term phase error processor (for example, short-term filter) 1904in phase error processor 314 establishes a phase tracking bandwidth ofthe phase and frequency locked loop. Long-term phase processor (forexample, filter) 1906 establishes a frequency tracking bandwidth of thephase and frequency locked loop. Short-term filter 1904 responds morequickly to phase changes in serial data signal 104 than does long-termfilter 1906. As a result, short-term absences of serial data signal 104(caused by signal drop-outs and the like, for example) can cause thephase and frequency locked loop to lose track of the serial data signalphase, since short-term filter 1904 is responsive to such short-termsignal losses. Therefore, after such signal losses, the phase andfrequency locked loop must re-acquire the serial data signal phase so asto re-establish a phase locked condition.

On the other hand, such short-term signal absences have less of anadverse affect on long-term filter 1906. Therefore, once the phase andfrequency locked loop begins rotating the sampling signal phase at aninitial rate to frequency synchronize the sampling and serial datasignals 208 and 104, the phase and frequency locked loop tends tocontinue rotating the sampling signal phase at the same initial rateduring the short-term signal losses. Therefore, when serial data signal104 returns after such a signal loss, sampling signal 208 tends to stillbe frequency synchronized with serial data signal 104 (assuming theserial data signal frequency does not change substantially during thesignal loss). Thus, the phase and frequency locked loop need onlyre-establish the phase locked condition mentioned above, since the loopis still frequency synchronized with serial data signal 104. Thisadvantageously reduces the time required to re-acquire the phase lockedcondition.

FIG. 20A is a flow chart of an example method 2000 of tracking thefrequency of serial data signal 104 using phase rotation according tothe present invention.

Method 2000 expands on steps 720 and 730 of method 700 described abovein connection with FIG. 7A. Step 720 includes steps 2002, 2004, and2006. At step 2002, short-term phase error processor 2002 derivesshort-term phase adjust signal 1910 by, for example, short-termfiltering phase error signal 350.

At next step 2004, frequency estimator 1906 estimates the frequencyoffset Δω between sampling signal 208 and serial data signal 104.Frequency estimator 1906 derives the frequency offset estimate by, forexample, long-term filtering of phase error 350.

At a next step 2006, rotate command generator 1908 generates phaserotate commands (for example, commands 354 a and/or 354 b) to compensatefor both the short-term phase offset and the frequency offset Δω.

Next step 730 includes a step 2010. At step 2010, phase control signalrotator 304 rotates digital phase control signals 340 andcorrespondingly interpolated phase φ_(I) of sampling signal 208 inresponse to phase rotate commands (such as commands 354 a and 354 b),such that sampling signal 208 and serial data signal 104 become phasealigned and frequency synchronized with one another.

The term “frequency synchronized” means sampling frequency ω_(s) andserial data signal frequency ω_(d) are matched to one another, such thatdata sample times t_(d) established by the frequency of sampling signal208, and coinciding with optimum symbol sample times t₀, do not “drift”relative to the symbol sample times t₀, over time. For this to be thecase in the present invention, sampling frequency ω_(s) and serial datasignal frequency ω_(d) need to be related to one another, but notnecessarily equal to one another, such that the frequencies aresynchronized. For example, frequencies ω_(s) and ω_(d) are consideredsynchronized to one another when ω_(d)=n·ω_(s), where n is an integergreater than one.

To decrease frequency ω_(s) relative to reference frequency ω_(r) (andserial data frequency ω_(d)) in the present invention, sampling signalphase φ_(I) is rotated in the clockwise direction (that is, in thedirection of increasing phase) at the necessary rate. On the other hand,to increase frequency ω_(s), phase φ_(I) is rotated in thecounter-clockwise direction (that is, in the direction of decreasingphase) at the necessary rate (for example, at a rate equal to thefrequency offset Δω). For example, with reference again to example phasering 1600 of FIG. 16A, the present invention rotates phasor or phasevalue 1620 in the clockwise direction around phase ring 1600 to decreasefrequency ω_(s) by an amount equal to the rate of rotation. On the otherhand, the present invention rotates phasor or phase value 1620 in thecounter-clockwise direction around phase ring 1600 to increase frequencyω_(s) by an amount equal to the rate of rotation.

FIG. 20B is a flow chart of an example method 2015 expanding on rotatingstep 2010 of method 2000. A step 2020 is initiated when the frequencyω_(s) of sampling signal 208 is greater than the frequency ω_(d) ofserial data signal 104 (i.e., when ω_(s)>ω_(d)) whereby step 2020decreases the frequency of the sampling signal, and correspondingly,reduces frequency offset Δω.

On the other hand, a step 2025 is initiated when the frequency ofsampling signal 208 is less than the frequency of serial data signal 104(i.e., when ω_(s)<ω_(d)), whereby step 2025 increases the frequency ofthe sampling signal, and correspondingly, reduces frequency offset Δω.

Example timing recovery systems 202 and 1810 include control signalrotator 304 for rotating phase control signals 340, and correspondingly,the interpolated phase of sampling signals 208 and 344. However, thepresent invention is not limited to such embodiments. For example, FIG.20C is a block diagram of an example timing recovery system 1845 forsynchronizing sampling and serial data signal frequencies, without usinga control signal rotator. Instead, timing recovery system 1845 includesa phase interpolator 306′ and a controller 1850. Phase interpolator 306′can be any known phase interpolator capable of adjusting theinterpolated phase of sampling signal 208 in response to an interpolatorcontrol signal 340′ (which may be a signal set 340′) compatible with thephase interpolator. For example, in a conventional configuration ofphase interpolator 306′ including multiplexer selectors for selectingbetween different signal phase to produce interpolated phases ofsampling signal 208, control signal set 340′ may include multiplexerselect signals, and so on.

Timing recovery system 1845 also includes phase detector 312 coupled toa phase error processor 314′. Phase error processor 314′ includes afrequency estimator to derive a frequency estimate (that is, a frequencymeasurement) of the frequency offset between sampling signal 208 andserial data signal 206, as described above, for example. In analternative arrangement, phase detector 312 and phase error processor314′ are combined into a single logic block for detecting the frequencyoffset. Phase error processor 314′ provides control signal 340′,indicative of the frequency offset, to phase interpolator 306′. Inresponse to control signal(s) 340′, phase interpolator 306′ rotates theinterpolated phase of sampling signal 208 to reduce the frequency offsetbetween the sampling signal and serial data signal 104.

FIG. 20D is a flow chart of a high level example method 2000′ offrequency synchronizing and phase-aligning sampling signal 208 to serialdata signal 104. Method 2000′ is similar to method 2000, and can beimplemented by either of timing control systems 202 and 1845. Method2000′ includes a step 720′ similar to step 720 of method 2000. However,step 720′ includes a generalized sub-step 2006′. In step 2006′,controller 1820/1850 (of timing system 202/1845) manipulates phasecontrol signals 340/340′, applied to phase interpolator 306/306′, inresponse to the detected phase and frequency offsets, so as to controlthe interpolated phase of sampling signal 208. For example, controller1820 rotates phase control signals 340 using rotator 304 (rotatingcontrol signals 340 was previously described as part of step 730/2010 inFIG. 20A, but is moved into step 720′ of method 2000′).

On the other hand, controller 1850 can manipulate phase control signals340′ in other ways, as would be apparent to one of ordinary skill in theart after reading the description provided herein. For example,controller 1850 can modify the values (for example, logic “1” or “0”) ofvarious ones of the phase control signals in accordance with the phaseand frequency offset, instead of rotating the phase control signals, soas to correspondingly rotate the interpolated phase of sampling signal208. Phase error processor 314′ can include formatting/generating logicto generate and/or manipulate phase control signals 340′ such that thephase control signals are compatible with phase interpolator 306′.

A next step 730′ is similar to step 730 of method 2000. In step 730′,interpolator 306/306′ rotates the interpolated phase of sampling signal208 in response to phase control signals 340/340′. Step 730′ is similarto step 730 to the extent phase interpolator 306 rotates theinterpolated phase of sampling signal 208 in response to phase controlsignals 340. However, step 730′ does not include rotating phase controlsignals 340, since this step is subsumed by previous step 720′ in method2000′, as described above.

FIG. 20E is a flow chart of a high level example method 2060 offrequency synchronizing sampling signal 208 to serial data signal 104.

An initial step 2064 includes deriving sampling signal 208 having aninterpolated phase (using phase interpolator 306/306′, for example).

A next step 2070 includes estimating a frequency offset between samplingsignal 208 and serial data signal 104 (using phase error processor314/314′, for example).

A next step 2075 includes rotating the interpolated phase of samplingsignal 208 at a rate corresponding to the frequency offset, so as toreduce the frequency offset.

IX. High-Speed Serial Transceiver

FIG. 21 is an illustration of an example multiple channel communicationdevice 2100 constructed on an integrated circuit (IC) chip 2102,according to an embodiment of the present invention. Communicationdevice 2100 is a multiple channel (that is, multi-channel) transceiver,including multiple receivers and multiple transmitters, as describedbelow. Each of the serial data signals is associated with a differentchannel. Communication device 2100 receives multiple analog serial datasignals 2104 a, 2104 b, 2104 c, and 2104 d (collectively referred to asmultiple serial data signals 2104). Communication device 2100 includesmultiple receive-lanes 2106 a, 2106 b, 2106 c, and 2106 d (collectivelyreferred to as multiple receive-lanes 2106, and each being associatedwith a receiver/receive-channel of communication device 2100). Each ofreceive-lanes 2106 receives a corresponding one of multiple serial datasignals 2104, as depicted in FIG. 21. Each of receive-lanes 2106processes the corresponding one of serial data signals 2104 to produce acorresponding one of multiple digital data streams 2108 a, 2108 b, 2108c, and 2108 d (collectively referred to as digital data streams 2108).Receive-lanes 2106 provide data streams 2108 to a digital data sampleprocessor 2112. Communication device 2100 is referred to as a multiplereceiver or multi-channel communication device because of the multiplereceive-lanes 2106 and associated circuits, described below.

Communication device 2100 includes a master timing generator 2114 forgenerating a master timing signal 2116. Master timing generator 2114 caninclude a reference oscillator and a PLL, such as reference oscillator330 and PLL 332, described above in connection with FIG. 3. Mastertiming generator 2114 provides master timing signal 2116 to each of themultiple receive-lanes 2106. In one arrangement, to minimize signalcrosstalk and interference in the present invention, master timingsignal 2116 includes a pair of differential (that is, complementary)clock signals/waves routed to each of receive-lanes 2106 over a pair ofclock lines.

Communication device 2100 also includes multiple transmit-lanes 2130 a,2130 b, 2130 c, and 2130 d (collectively referred to as multipletransmit-lanes 2130). Data sample processor 2112 provides multipletransmit data streams 2134 a, 2134 b, 2134 c, and 2134 d (collectivelyreferred to as multiple transmit digital data streams 2134) tocorresponding ones of transmit-lanes 2130, as depicted in FIG. 21.Master timing generator 2114 provides master timing signal 2116 to eachof the multiple transmit-lanes 2130. Transmit-lanes 2130 each transmit acorresponding one of multiple analog serial data signals 2140 a, 2140 b,2140 c, and 2140 d (collectively referred to as multiple transmit analogserial data signals 2140). In alternative embodiments, communicationdevice 2100 may include more or fewer receive-lanes 2106 andtransmit-lanes 2130.

Communication device 2100 can include more or less than four receiverand/or transmit lanes in other embodiments.

FIG. 22 is an illustration of receive-lane 2106 a, according to amultiple data path per receive-lane embodiment of the present invention.In an embodiment, exemplary receive-lane 2106 a is substantiallyidentical to the other receive-lanes 2106 b-d, therefore the followingdescription of receive-lane 2106 a shall suffice for the others.Receive-lane 2106 a includes a data module 2204, a phase module 2206,and a sampling signal generator 2208. Also depicted in FIG. 22 isdigital data sample processor 2112. As depicted, processor 2112 includesa data demultiplexer module 2210 a and an interpolator control module2212 a, both corresponding to receive-lane 2106 a. Processor 2112provides interpolator phase control signals 2214 a, including a firstphase control signal set 2214 a ₁ and a second phase control signal set2214 a ₂, to sampling signal generator 2208.

Sampling signal generator 2208 derives a plurality of timing signalsrequired to operate receive-lane 2106 a from master timing signal 2116,as described below. An advantage of deriving such timing signals locallywithin receive-lane 2106 a, is to reduce signal/clock cross-talk andinterference across IC chip 2102, and to reduce the number of signaltraces or tracks distributed across the IC chip.

Sampling signal generator 2208 includes a first signal set generator2220. First signal set generator 2220 derives a set of reference signals2222 having different predetermined phases from master timing signal2116. Signal set 2222 can be the same as or similar to reference signalset 206 described above in connection with FIG. 15A, for example. Signalset generator 2220 provides signal set 2222 to a phase interpolatormodule 2224.

Phase interpolator module 2224 receives signal set 2222 and phasecontrol signals 2214 a from processor 2220. In the embodiment depictedin FIG. 22, phase interpolator module 2224 includes first and secondphase interpolators 2226 ₁ and 2226 ₂. Each of phase interpolators 2226₁ and 2226 ₂ receives signal set 2222, together with a respective one ofphase control signal sets 2214 a ₁, and 2214 a ₂ included in phasecontrol signals 2214 a. In response to these signal inputs, phaseinterpolators 2226 ₁ and 2226 ₂ respectively derive interpolated timingsignals 2230 ₁ and 2230 ₂ (collectively referred to as interpolatedtiming signals 2230). Phase interpolator module 2224 providesinterpolated timing signals 2230 to a second signal set generator 2234.

Second signal set generator 2234 derives multiple time-staggered dataand phase sampling signals 2238 from interpolated timing signals 2230.Therefore, time-staggered data and phase sampling signals 2238 each hasan interpolated phase corresponding to the interpolated phase of timingsignals 2230. Time-staggered data and phase sampling signals 2238include time-staggered data sampling signals d0, d1, d2, and d3, andtime-staggered phase sampling signals x0, x1, x2, and x3. Signal setgenerator 2234 generates the multiple time-staggered data and samplingsignals 2238 such that data sampling signal d0 and phase sampling signalx0 are paired with one another, data sampling signal d1 and phasesampling signal x1 are paired with one another, and so on.

In the example embodiment depicted in FIG. 22, master timing signal 2116has a frequency equal to the symbol frequency B (that is, baud rate B)of serial data signal 2104 a. First signal set generator 2220 includesfour-phase clock generator divide-by-two divider circuits, such thatgenerator 2220 generates four signals in signal set 2222, each at afrequency B/2. Thus, phase interpolators 2226 ₁ and 2226 ₂ producerespective interpolated timing signals 2230 ₁ and 2230 ₂ each at acorresponding frequency of B/2. In an embodiment, timing signals 2230 ₁,and 2230 ₂ have respective phases offset from each other by 90°. Secondsignal set generator 2234 includes eight-phase clock generatordivide-by-two divider circuits, to produce each of the eight data andsampling signals 2238 at a frequency B/4.

Data module 2204 includes multiple parallel data paths 2242 ₀, 2242 ₁,2242 ₂, and 2242 ₃ (collectively referred to as data paths 2242). Eachof the data paths 2242 ₀, 2242 ₁, 2242 ₂, and 2242 ₃ receives serialdata signal 2104 a. Each of data paths 2242 ₀, 2242 ₁, 2242 ₂, and 2242₃ samples serial data signal 2104 a according to a corresponding one oftime-staggered data sampling signals d0, d1, d2, and d3, therebyproducing corresponding multiple time-staggered data sample streams 2244₀, 2244 ₁, 2244 ₂, and 2244 ₃ (collectively referred to as multipletime-staggered data sample streams 2244), as depicted in FIG. 22.Therefore, multiple data paths 2242 provide multiple data streams 2244to processor 2112. The use of multiple parallel data sampling pathswithin a receive-lane in the present invention, as depicted in FIG. 22,for example, facilitates processing of high frequency serial datasignals, such as a serial data signal having a multi-gigabit symbolrate, because each of the parallel data paths can sample the serial datasignal at a rate below the multi-gigabit symbol rate, as will be furtherdescribed below.

Phase module 2206 includes multiple phase paths 2250 ₀, 2250 ₁, 2250 ₂,and 2250 ₃ (collectively referred to as multiple phase paths 2250). Eachof the phase paths in multiple phase paths 2250 samples serial datasignal 2104 a according to a corresponding one of time-staggered phasesampling signals x0, x1, x2, and x3, as depicted in FIG. 22, therebyproducing multiple time-staggered phase sample streams 2252 ₀, 2252 ₁,2252 ₂, and 2252 ₃ (collectively referred to as phase sample streams2252). Data streams 2244 and phase streams 2252 collectively form datastream 2108 a depicted in FIG. 21. In alternative embodiments,receive-lane 2106 a can include more or fewer data and phase paths 2242and 2250. Also, different ones of receive-lanes 2106 can have differentnumbers of data paths and different numbers of phase paths. Also,sampling signal generator 2208 in each receive-lane can derive more orless time-staggered data and phase sampling signals according to thenumber of parallel data and phase paths in the receive-lane. Samplingsignal generator 2208 can include less or more phase interpolators, asthe need arises to generate more or less timing and sampling signals inthe receive-lane.

In an embodiment, each of data paths 2242 and phase paths 2250 aresubstantially identical, and therefore, the following description ofexemplary data path 2242 ₀ shall suffice for the other data and phasepaths in such an embodiment. Data path 2242 ₀ includes a sampler 2260,an equalizer 2262 following sampler 2260, and a quantizer 2264 followingthe equalizer. Sampler 2260 samples analog serial data signal 2104 a atsample times established by data sampling signal d0, to produce asampled analog data signal 2270 representative of serial data signal2104 a. Equalizer 2262 equalizes sampled analog data signal 2270 toproduce an equalized, sampled analog data signal 2272. Thus, equalizer2262 reduces inter-symbol interference present in serial data signal2104 a. Quantizer 2264 quantizes analog samples of sampled analog signal2272 into corresponding, quantized digital data samples. Quantizer 2264provides signal 2244 ₀, including the quantized digital data samples, toprocessor 2112. Exemplary further details of data and phase pathsincluding equalizers are provided in U.S. Non-Provisional applicationSer. No. 09/844,283, filed Apr. 30, 2001, now U.S. Pat. No. 7,286,597,entitled “Methods and Systems for Adaptive Receiver Equalization,”incorporated herein by reference in its entirety.

Data demultiplexer module 2210 a receives multiple time-staggered (thatis, time-skewed) data streams 2244. Data demultiplexer module 2210 atime-deskews and then demultiplexes/deserializes multiple time-staggereddata streams 2244, to produce a demultiplexed data sample stream 2280 arepresentative of serial data signal 2104 a. Demultiplexed data samplestream 2280 a includes quantized digital data samples arranged in aparallel word format. Therefore, data demultiplexer module 2210 a can beconsidered a deserializer or serial-to-parallel converter module.

Interpolator control module 2212 a receives multiple data streams 2244from data module 2204 and multiple phase streams 2252 from phase module2206. Interpolator control module 2212 a detects phase and frequencyoffsets between multiple time-staggered data sampling signals d0-d3 andserial data signal 2104 a. Interpolator control module 2212 a derivesinterpolator phase control signals 2214 a in response to the detectedphase and frequency offsets, as described above. In response to phasecontrol signals 2214, phase interpolator module 2224 rotates theinterpolated phase of timing signals 2230, and correspondingly oftime-staggered data and phase sampling signals 2238, to compensate forthe detected phase offset and at a rate corresponding to the detectedfrequency offset, as described above. In this manner, interpolatorcontrol module 2212 a causes time-staggered data sampling signals d0-d3to be phase-aligned and frequency-synchronized with serial data signal2104 a.

FIG. 23 is an illustration of various example signal waveforms (b)-(j)from receive-lane 2106 a depicted in FIG. 22.

Waveform (a) represents a clock wave 2302 having a frequency B=1/Tcorresponding to a symbol rate of serial data signal 2104 a.

Waveform (b) represents serial data signal 2104 a, including consecutiveNRZ symbols 2304, each having a symbol period T.

Waveforms (c)-(j) respectively represent time-staggered data and phasesampling signals d0, x0, d1, x1, d2, x2, d3, and x3. As depicted in FIG.23, each sampling signal (for example, d0) is offset in time (that is,time-staggered or time-skewed) from the next sampling signal (forexample, x0) by a half symbol period (that is, by a time offset=T/2).Therefore, consecutive data sampling signals (for example, d0, d1, andd1, d2) are time-staggered by a symbol period T. Each of the samplingsignals d0-d3, and x0-x3 has a sampling signal period=4·T (that is, asampling signal frequency of one-quarter the symbol rate of serial datasignal 2104 a). As a result, in each sampling signal period 4·T, datasampling signals d0-d3 cause data paths 2242 ₀₋₃ to collectively samplefour consecutive symbols of serial data signal 2104 a, such that eachdata path samples a different one of the four consecutive symbols. In anexample implementation of the present invention, serial data signal 2104a has a symbol rate=3.125 GHz, and each of sampling signals d0-x3 has asampling signal rate=781.25 MHz.

FIG. 24 is a phase circle 2400 representing the evenly spaced phases ofsampling signals d0-x3 depicted in FIG. 23. A phase rotation of 360°corresponds to a sampling signal period of 4·T.

FIG. 25 is a block diagram of data demultiplexer module 2210 a,according to an embodiment of the present invention. Data demultiplexermodule 2210 a includes a data deskewer 2502 followed by a datademultiplexer/deserialize 2504. Data deskewer 2502 receives multipletime-staggered data sample streams 2244 and multiple data samplingsignals d0-d3. Data deskewer 2502 time-deskews (that is, removes thetime offset between) multiple data sample streams 2244, and presentscorresponding deskewed data sample streams 2510 ₀₋₃ to demultiplexer2504. For example, in each data sampling period, data deskewer 2502receives four time-staggered symbol samples from data sample streams2244, collectively. Data deskewer 2502 time-deskews the four datasamples, and presents four corresponding deskewed data samples todemultiplexer 2504 (in multiple deskewed data streams 2510 ₀₋₃).

Data demultiplexer 2504 deserializes/demultiplexes the deskewed datasample streams 2510 ₀₋₃ to produce deserialized/demultiplexed datasample stream 2280 a. Demultiplexer 2504 includes a set, such as five,four-bit registers 2510 ₀₋₄, for example. During five consecutive datasampling periods, data demultiplexer module 2210 a consecutivelytransfers five sets of four deskewed data samples from deskewer 2502(that is, from data sample streams 2510 ₀₋₃) into correspondingconsecutive ones of the five four-bit registers 2512 ₀₋₄. Thus, twentyserialized data samples are transferred to registers 2512 ₀₋₄ indemultiplexer 2504. Demultiplexer 2504 constructs a twenty-bit wideparallel word including the twenty serialized data samples mentionedabove. Demultiplexer 2504 outputs the twenty-bit parallel wordrepresentative of the twenty serialized data samples in demultiplexeddata sample stream 2280 a. Demultiplexer 2504 can transfer thetwenty-bits as two ten-bit parallel words, for example.

FIG. 26 is a block diagram of interpolator control module 2212 a,according to embodiment of the present invention. Interpolator controlmodule 2212 a receives the multiple data streams 2244 and the multiplephase streams 2252. Interpolator control module 2212 a includes phasedetector 2212, phase error processor 2214, and a phase control signalrotator 2604. Phase control signal rotator 2604 includes a first phasecontrol signal rotator 2204 ₁ and a second phase control signal rotator2204 ₂ to correspondingly produce first and second phase control signalsets 2214 a ₁ and 2214 a ₂ of phase controls signals 2214 a. Otherembodiments of interpolator control module 2212 a are possible, as wouldbe apparent to one of ordinary skill in the relevant art(s). Forexample, and as described above in connection with FIG. 20C, theinterpolator control module is not limited to an embodiment including acontrol signal rotator.

Multiple data paths 2242, multiple phase paths 2250, phase detector2212, phase error processor 2214, and the signal rotators of phasecontrol signal rotator 2604, operate together in a manner consistentwith the description of the same or similar elements describedpreviously in connection with timing recovery module 202, for example.Therefore, receive-lane 2106 a includes a timing recovery system/module(such as timing recovery module 202) associated with the receive-lane,to phase and frequency track serial data signal 2104 a. In other words,the timing recovery module associated with receive-lane 2106 a adjuststhe interpolated phases of time-staggered data sampling signals d0-d3such that each of the sampling signals d0-d3 causes the correspondingone of data paths 2242 to optimally sample consecutive symbols in serialdata signal 2104 a. In accordance with the operation of the timingrecovery module associated with receive-lane 2106 a, sampling signal d0causes data path 2242 ₀ to sample a mid-point of a first symbol ofserial data signal 2104 a, sampling signal d1 causes data path 2242 ₁,to sample a mid-point of a next symbol of serial data signal 2104 a, andso on.

In addition, the timing recovery module associated with receive-lane2106 a frequency synchronizes data sampling signals d0-d3 with serialdata signal 2104 a. In other words, the timing recovery moduleassociated with receive-lane 2106 causes the interpolated phases of datasampling signals d0-d3 to rotate at a rate that synchronizes a commonfrequency, f_(s), of sampling signals d0-d3 to the symbol baud rate B ofserial data signal 2104 a. In the example embodiment depicted in FIG.21, the timing recovery module associated with receive-lane 2106 arotates the interpolated phases of data sampling signals d0-d3 at a ratesuch that the common sampling frequency f_(s)=B/4.

FIG. 27 is a block diagram of processor 2112, according to an embodimentof the present invention. Processor 2112 includes multiple datademultiplexer modules 2210 a, 2210 b, 2210 c, and 2210 d (collectivelyreferred to as data demultiplexer modules 2210), each corresponding tothe data sample streams (that is, data streams 2244) of one of datastreams 2108 a-2108 d. Processor 2112 also includes multipleinterpolator control modules 2212 a, 2212 b, 2212 c, and 2212 d(collectively referred to as interpolator control modules 2210), eachcorresponding to one of data streams 2108 a, 2108 b, 2108 c, and 2108 d.In other words, processor 2112 includes a data demultiplexer module andan interpolator control module for each receive channel of communicationdevice 2100.

Therefore, each of the receive-lane is associated with a separate timingrecovery module (such as timing recovery module 202), wherein eachtiming recovery module operates independently of each other timingmodule. This means the timing recovery module associated withreceive-lane 2106 a tracks a phase and a frequency of serial data signal2104 a, while the timing recovery module associated with receive-lane2106 b can track a different phase and a different frequency of serialdata signal 2104 b, and so on. For example, the interpolated phases ofthe sampling signals (d0-d3) associated with receive-lane 2106 a can berotated independently of and at a rate different from the interpolatedphases of the sampling signals associated with the other receive-lanes2106 b-c.

FIG. 28 is a block diagram of a communication device 2800, correspondingto communication device 2100, according to another embodiment of thepresent invention. Unlike the communication device embodiment depictedin FIG. 22, communication device 2800 does not include multiple parallelsampling paths within a receive-lane of the communication device, aswill be described below.

Communication device 2800 is constructed on an IC chip 2802.Communication device 2800 includes multiple receive-lanes 2804 a-n.Receive-lane 2804 a includes a sampling signal generator 2806 a, a datapath 2808 a, and a phase path 2810 a. An interpolator control module2812 a is included as part of a digital data processor, not shown.Sampling signal generator 2806 a includes signal set generator 2220 (asdescribed in connection with FIG. 22, for example), and a phaseinterpolator 2814. Interpolator control module 2812 a includes phasedetector 2212, phase error processor 2214, and phase control signalrotator 2204, as described previously. Phase interpolator 2814 providesinterpolated sampling signals 2815 ₁ and 2815 ₂ to respective data andphase paths 2808 a and 2810 a. Data path 2808 a and phase path 2810 acan include the same elements as are included in data path 2242 ₀,described in connection with FIG. 22. If this is the case, then datapath 2808 a and phase path 2810 a each provide serial, quantized,digital data samples (2816 ₁ and 2816 ₂, respectively) to the digitaldata processor (not shown). Alternatively, a datademultiplexer/deserializer can be added to each of data path 2802 a andphase path 2810 a, after quantizer 2264 in each path. Such a datademultiplexer after quantizer 2264 supplies demultiplexed data samples(in parallel word format) to the digital data processor.

FIG. 29 is a flow chart of an example method of processing a serial datasignal in multiple parallel data paths, using receive-lane 2106 adepicted in FIG. 22, for example. An initial step 2902 includesgenerating a master timing signal (for example, using master timinggenerator 2114).

A next step 2904, includes generating multiple time-staggered samplingsignals (such as signals d0-d3) based on the master timing signal.

A next step 2906 includes sampling a received, analog serial data signal(such as serial data signal 2104 a) in accordance with each of themultiple time-staggered sampling signals (for example, d0-d3), therebyproducing multiple time-staggered data sample streams (such as datasample streams 2244).

A next step 2908 includes time-deskewing the multiple time-staggereddata streams (for example, using deskewer 2502).

A next step 2910 includes demultiplexing multiple time-deskewed datastreams produced in step 2908 (using, for example, demultiplexer 2504depicted in FIG. 25).

FIG. 30 is a flow chart of an example method 3000 of frequencysynchronizing multiple data sampling signals (channels) to correspondingones of multiple serial data signals using communication device 2100.Method 3000 can be implemented using the communication deviceembodiments depicted in both FIGS. 22 and 28.

An initial step 3002 includes generating a master timing signal (usingmaster timing generator 2114, for example).

A next step 3004 includes deriving multiple sampling signals (such as asampling signal d0 in receive-lane 2106 a, sampling signal d0 inreceived-lane 2106 b, and sampling signal d0 in received-lane 2106 c)based on the master timing signal (for example, master timing signal2116). Each of the multiple sampling signals is associated with one ofmultiple serial data signals (for example, sampling signal d0 inreceive-lane 2106 a is associated with serial data signal 2104 a,sampling signal d0 in receive-lane 2106 b is associated with serial datasignal 2104 b, and so on). Each of the sampling signals has aninterpolated phase.

A next step 3006 includes sampling and quantizing each of the multipleserial data signals (2104 a, 2104 b, and so on) according to theassociated one of the sampling signals (for example, sampling signal d0in receive-lane 2104 a, and sampling signal d0 in receive-lane 2104 b,and so on).

A next step 3008 includes rotating the interpolated phase of eachsampling signal at a rate corresponding to a frequency offset betweenthe sampling signal and the serial data signal associated with thesampling signal (such as between sampling signal d0 in receive-lane 2106a and serial data signal 2104 a), whereby each sampling signal isfrequency synchronized with each associated serial data signal.

X. Example Transceiver Use

In an embodiment, the present invention is implemented as a signalrouter. A signal router can be used to route one or more informationsignals between a plurality of components.

FIG. 31 is an illustration of an example use of a transceiver of thepresent invention. The transceiver of the present invention is used inan example signal router 3100, including a front panel 3102, a backplane 3104 and one or more interfacing circuit boards 3106. Front panel3102 typically includes a plurality of connectors or “jacks,” to whichexternal devices, such as computers, servers, terminals, communicationsdevices, other routers, and the like, can be coupled. The router 3100receives and transmits (i.e., routes) signals between the externaldevices.

Each interfacing circuit board 3106 includes a finite number ofconnections to the front panel 3102 for receiving and/or transmittingsignals from/to external devices. Additional interfacing circuit boards3106 can be utilized to accommodate additional external devices. Thebackplane 3104 permits the router 3100 to route signals between multipleinterfacing circuit boards 3106. In other words, the backplane 3104permits the router 3100 to route signals between external devices thatare coupled to different interfacing circuit boards 3106.

Interfacing circuit boards 3106 can include a variety of digital and/oranalog components. When multiple interfacing circuit boards 3106 areutilized, two or more of them can be similar and/or dissimilar. Theinterfacing circuit boards 3106 illustrated in FIG. 31 are provided forillustrative purposes only. Based on the description herein, one skilledin the relevant art(s) will understand that additional and/oralternative components/features can be provided with the interfacingcircuit boards 3106.

Example interfacing circuit board 3106 is now described. Interfacingcircuit board 3106A optionally includes one or more interface components3108 that receive and/or buffer one or more signals received fromexternal devices through the front panel 3102. In the illustratedexample, the interface component 3108 receives an optical signal 3109from the front panel 3102. Accordingly, in this embodiment, interfacingcomponent 3108 includes one or more optical converters that convert theoptical signal 3109 to an electrical analog data signal, illustratedhere as an analog serial data signal 3112. Additionally, oralternatively, interfacing component 3108 sends and/or receives one ormore other analog data signals 3114A-n to/from other external devicesthrough the front panel 3102. Additionally, or alternatively,interfacing component 3108 sends and/or receives one or more of thesignals 3114A-n to/from somewhere other than the front panel 3102.

The serial analog data signal 3112 is provided from the interfacingcomponent 3108 to a transceiver 3110, which can be implemented as one ormore of transceivers 2100 (FIG. 21), for example. Transceiver 3110permits the router 3100 to both receiver and transmit analog serial data3112 from and to external devices.

Within the transceiver 3110, a receiver portion 3111 (includingreceive-lanes 2106, master timing generator 2114, and digital datasample processor 2112, for example) converts the serial analog datasignal 3112 to one or more digital data signals, illustrated here asparallel digital data signals 3116.

The parallel digital data signals 3116 are optionally provided to aswitch fabric 3118, which can be a programmable switching fabric. Theoptional switching fabric 3118 provides any of a variety offunctionalities.

The optional switching fabric 3118 outputs parallel digital data signals3120 to second transceiver 3122, which can be implemented as one or moreof transceivers 2100 (FIG. 21), for example. A transmitter portion 3123(including transmit-lanes 2130 and digital data sample processor 2112,for example) within the transceiver 3122 converts the parallel digitaldata signals 3120 to serial analog data signals 3124 and transmits themacross the back plane 3104 to other interface circuit boards 3106 n,and/or back to interface circuit board 3106A.

A receiver portion 3111 within the transceiver 3122 receives analog datasignals 3124 from the back plane 3104 and converts them to paralleldigital data signals 3120. The parallel digital data signals 3120 areprovided to the switch fabric 3118, which provides any of a variety offunctionalities. The switch fabric 3118 outputs parallel digital datasignals 3116 to a transmitter 3123 within the transceiver 3110, whichconverts them to analog data signals for transmission to an externaldevices, possibly through the interface component 3108 and the frontpanel 3102.

Additional interface circuit boards 3106 n operate in a similar fashion.Alternatively, one or more of the interface circuit boards 3106A-n areconfigured with more or less than the functionality described above. Forexample, in an embodiment, one or more of the interface circuit boards3106A-n are configured to receive analog data signals from the frontpanel 3102 and to provide them to the back plane 3104, but not toreceive analog data signals 3124 from the back plane 3104.Alternatively, one or more of the interface circuit boards 3106A-n areconfigured to analog data signals 3124 from the back plane 3104 andprovide them to the front panel, but not to receive analog data signalsfrom the front panel 3102.

XI. Further Phase Interpolator Implementations

As described herein, embodiments of the present invention include aphase interpolator 306 that may be implemented in the manner describedabove with reference to FIGS. 8-14B. However, other implementations maybe employed for phase interpolators 306, 306′, 2226, and 2814. Twoalternative example implementations are illustrated in FIGS. 32 and 33.

FIG. 32 is a block diagram of a phase interpolator implementation 3200.Implementation 3200 includes four reference stages 3202 a-d. Like thephase interpolator 801 implementations of FIGS. 8-14B, reference stages3202 a-d receive reference signals 820 a-d, respectively. Further,reference stages 802 a-d also receive control signals 822 a-d,respectively.

Like the phase interpolator implementations described above withreference to FIGS. 8-14B, each reference stage 3202 generates acomponent signal 824 from its corresponding reference signal 820according to a scaling factor that is the ratio of its component signal824 magnitude to its corresponding reference signal 820 magnitude. Thisscaling factor is determined by corresponding control signal 822,through the use of variable gain amplifiers (VGAs) 3204. Each componentsignal 824 is combined (e.g., summed) at combining node 804 to produceoutput signal 826, having an interpolated phase.

As shown in FIG. 32, each reference stage 3202 includes a VGA 3204 thatreceives a corresponding reference signal 820 and a correspondingcontrol signal 822. For example, reference stage 3202 a receivesreference signal 820 a and control signal 822 a. Each VGA 3204 has again that determined by the value of it corresponding control signal 822according to a predetermined relationship. In one such relationship,gain increases as the control signal 822 increases. In an alternativerelationship, gain decreases as the control signal 822 increases.

The scaling factor of each reference stage 3202 is determined by thegain of its VGA 3204. In particular, for example, as the gain increases,so does the corresponding reference stage 3202 scaling factor.

FIG. 33 is a block diagram of a phase interpolator implementation 3300.Like implementation 3200, implementation 3300 includes amplifiers.However, implementation 3300 provides adjustable scaling factors throughvariable resistance.

As shown in FIG. 33, implementation 3300 includes four reference stages3302 a-d that each include a constant gain amplifier 3304 that iscoupled to a variable resistance 3306. Like the phase interpolator 801implementations of FIGS. 8-14B, reference stages 3302 a-d receivereference signals 820 a-d, respectively. Further, reference stages 802a-d also receive control signals 822 a-d, respectively. For eachreference stage 3302, its amplifier 3304 receives the correspondingreference signal 820 and its variable resistance 3306 receives thecorresponding control signal 822.

Each reference stage 3302 generates a component signal 824 from itscorresponding reference signal 822 according to a scaling factor that isthe ratio of its component signal 824 magnitude to its correspondingreference signal 820 magnitude. This scaling factor is determined bycorresponding component signal 822, through the use of variableresistances 3306. Each component signal 824 is combined (e.g., summed)at combining node 804 to produce output signal 826, having theinterpolated phase.

The scaling factor of each reference stage 3302 is determined by thevalue of its variable resistance 3306. As shown in FIG. 33, eachvariable resistance 3306 receives a corresponding control signal 822.The value of each variable resistance 3306 is determined by the value ofits corresponding control signal 822 according to a predeterminedrelationship. In one such relationship, resistance decreases as thecontrol signal 822 increases. Alternatively, resistance increases as thecontrol signal 822 increases.

The scaling factor of each reference stage 3302 is determined by thevalue of its variable resistance 3306. In particular, as the resistanceincreases, the corresponding reference stage 3202 scaling factordecreases.

Each of the phase interpolators described above are responsive todigital phase control signals for controlling the interpolated phaseproduced by the interpolator. Thus, such phase interpolators can beadvantageously used in digital timing recovery systems implemented as“all” digital timing recovery systems including all digital controlloops. This can advantageously improve reliability in producing andoperating such timing recovery systems. However, it is to be understoodthat the present invention can also include phase interpolatorsresponsive to analog phase control signals for controlling theinterpolated phase. For example, the present invention can includereference stages (including VGAs, variable resistances, IDACs, and thelike) responsive to phase control signals, each having multiple analoglevels, to control the magnitudes of corresponding component signals,and thus, the interpolated phase.

Each of phase interpolators 306, 306′, 2226 ₁, 2226 ₂, and 2814,described above, can be implemented in many ways, as would be apparentto one of ordinary skill in the relevant art(s) after reading thedescription provided herein.

XII. Conclusion

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample only, and not limitation. For example, aspects of the presentinvention are described above in the context of a phase interpolatorhaving four reference stages. However, the present invention may includephase interpolators having any number of reference stages.

For instance, the present invention may include a three reference stagephase interpolator. In this embodiment, each reference stage receivesone of three reference signals that are offset in phase by 120 degrees.Alternatively, the present invention may include a two reference stagephase interpolator, each reference stage receiving one of two referencesignals having spaced phases.

Finally, it will be understood by those skilled in the art that variouschanges in form and details may be made therein without departing fromthe spirit and scope of the invention as defined in the appended claims.Thus, the breadth and scope of the present invention should not belimited by any of the above-described exemplary embodiments, but shouldbe defined only in accordance with the following claims and theirequivalents.

What is claimed is:
 1. A method for sampling a high data rate signal fora communication device, comprising: sampling a received serial datasignal using a plurality of time-staggered sampling signals to generatea plurality of time-staggered sampled signals; detecting time offsetsbetween the plurality of time-staggered sampled signals, the timeoffsets representing differences between respective time-staggeredsampling signals and the received serial data signal; and adjusting theplurality of time-staggered sampling signals based on the time offsets.2. The method of claim 1, further comprising: time-deskewing theplurality of time-staggered sampled signals to generate a plurality oftime-deskewed sampled signals.
 3. The method of claim 2, furthercomprising: generating a parallel signal from the plurality oftime-deskewed sampled signals, wherein the parallel signal includessamples of the received serial data signal.
 4. The method of claim 1,wherein adjusting the plurality of time-staggered sampling signalscomprises: adjusting the plurality of time-staggered sampling signalssuch that each of the plurality of time-staggered sampling, signals hasan interpolated phase.
 5. The method of claim 1, wherein adjusting theplurality of time-staggered sampling signals comprises: adjusting theplurality of time-staggered sampling signals based on a timing signal;generating a set of reference signals having different phases from thetiming signal; generating a second timing signal having an interpolatedphase from the set of reference signals; and adjusting the plurality oftime-staggered sampling signals based on the second timing signal,wherein each of the plurality of time-staggered sampling signals has aninterpolated phase related to the interpolated phase of the secondtiming signal.
 6. The method of claim 1, wherein the sampling of thereceived serial data signal comprises: sampling the received serial datasignal at a plurality of sampling times, each of the plurality ofsampling times corresponding to a respective one of the plurality oftime-staggered sampling signals.
 7. A communication device, comprising:a sampler that comprises a plurality of sampling paths, the plurality ofsampling paths being configured to sample a received serial data signalusing a corresponding one of a plurality of time-staggered samplingsignals, thereby generating a plurality of time-staggered sampledsignals; a time-deskewer coupled to the sampler, wherein thetime-deskewer is configured to remove time offsets between the pluralityof time-staggered sampled signals, the time offsets representingdifferences between respective time-staggered sampling signals and thereceived serial data signal; and a sampling signal generator coupled tothe time-deskewer, wherein the sampling signal generator is configuredto adjust the plurality of time-staggered sampling signals based on thetime offsets.
 8. The communication device of claim 7, wherein thetime-deskewer is configured to time-deskew the plurality oftime-staggered sampled signals to generate a plurality of time-deskewedsampled signals, wherein the plurality of time-deskewed sampling signalsare characterized as having approximately zero time offset relative toone another.
 9. The communication device of claim 8, further comprising:a deserializer configured to generate a parallel signal from theplurality of time-deskewed sampled signals, wherein the parallel signalincludes samples of the received serial data signal.
 10. Thecommunication device of claim 7, wherein the sampling signal generatoris further configured to generate the plurality of time-staggeredsampling signals based on a timing signal, and wherein the samplingsignal generator comprises: a signal generator configured to generate aset of reference signals having different phases from the timing signal;and a phase interpolator module configured to generate a second timingsignal having an interpolated phase from the set of reference signalsand to adjust the plurality of time-staggered sampling, signals from thesecond timing signal, wherein each of the plurality of time-staggeredsampling signals has an interpolated phase related to the interpolatedphase of the second timing signal.
 11. The communication device of claim7, wherein each of the plurality of sampling paths comprises: aquantizer configured to quantize a corresponding one of the plurality oftime-staggered sampled signals.
 12. The communication device of claim 7,wherein the sampling signal generator comprises: a first signal setgenerator, wherein the first signal set generator is configured togenerate a set of reference signals having different predeterminedphases; an interpolator module coupled to the first signal setgenerator, wherein the interpolator module is configured to generate aplurality of interpolated timing signals based on the set of referencesignals and a plurality of phase control signals; and a second signalset generator coupled to the interpolator module, wherein the secondsignal set generator is configured to adjust the plurality oftime-staggered sampling signals based on the plurality of interpolatedtiming signals.
 13. A sampling signal generator, comprising: a firstsignal set generator, wherein the first signal set generator isconfigured to generate a set of reference signals having differentpredetermined phases; an interpolator module coupled to the first signalset generator, wherein the interpolator module is configured to generatea plurality of interpolated timing signals based on the set of referencesignals and a plurality of phase control signals; and a second signalset generator coupled to the interpolator module, wherein the secondsignal set generator is configured to: generate a plurality oftime-staggered sampling signals based on the plurality of interpolatedtiming signals, and adjust the plurality of time-staggered samplingsignals based on a plurality of time offsets, the plurality of timeoffsets representing differences between respective time-staggeredsampling signals and a received serial data signal.
 14. The samplingsignal generator of claim 13, wherein the interpolator module is furtherconfigured to receive the plurality of phase control signals from aninterpolator control module coupled to the interpolator module.
 15. Thesampling signal generator of claim 13, wherein the second signal setgenerator is configured to adjust the plurality of time-staggeredsampling signals based on the plurality of interpolated timing signals.16. The sampling signal generator of claim 13, wherein the first signalset generator is further configured to: receive a master timing signal;and generate the set of reference signals based on the master timingsignal.
 17. A method for sampling a high data rate signal for acommunication device, comprising: generating a set of reference signalshaving different predetermined phases; generating a plurality ofinterpolated timing signals based on the set of reference signals and aplurality of phase control signals; generating a plurality oftime-staggered sampling signals based on the pluralty of interpolatedtiming signals; and adjusting the plurality of time-staggered samplingsignals based on a plurality of time offsets, the plurality of timeoffsets representing differences between respective time-staggeredsampling signals and a received serial data signal.
 18. The method ofclaim 17, further comprising: receiving the plurality of phase controlsignals from an interpolator control module.
 19. The method of claim 17,further comprising: adjusting the plurality of time-staggered samplingsignals based on the plurality of interpolated timing signals.
 20. Themethod of claim 17, further comprising: receiving a master timingsignal; and generating the set of reference signals based on the mastertiming signal.